Document Outline
- FEATURES
- APPLICATIONS
- GENERAL DESCRIPTION
- PIN CONFIGURATIONS
-
-
-
-
-
Precision 20 MHz CMOS Rail-to-Rail
Input/Output Operational Amplifiers
AD8616/AD8618
Rev. A
Information furnished by Analog Devices is believed to be accurate and reliable.
However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
or otherwise under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.326.8703
2004 Analog Devices, Inc. All rights reserved.
FEATURES
Low offset voltage: 65 V max
Single-supply operation: 2.7 V to 5.5 V
Low noise: 8 nV/Hz
Wide bandwidth: >20 MHz
Slew rate: 12 V/s
High output current: 150 mA
No phase reversal
Low input bias current: 1 pA
Low supply current: 2 mA
Unity gain stable
APPLICATIONS
Barcode scanners
Battery-powered instrumentation
Multipole filters
Sensors
ASIC input or output amplifier
Audio
Photodiode amplification
PIN CONFIGURATIONS
OUT A
1
IN A
2
+IN A
3
V
4
V+
8
OUT B
7
IN B
6
+IN B
5
AD8616
TOP VIEW
(Not to Scale)
04648-0-001
Figure 1. 8-Lead MSOP (RM-8)
OUT A
1
IN A
2
+IN A
3
V
4
V+
8
OUT B
7
IN B
6
+IN B
5
AD8616
TOP VIEW
(Not to Scale)
04648-0-002
Figure 2. 8-Lead SOIC (R-8)
OUT A
IN A
+IN A
V+
+IN B
IN B
OUT B
IN D
+IN D
V
OUT D
IN C
OUT C
+IN C
14
8
1
7
AD8618
04648-0-048
Figure 3. 14-Lead TSSOP (RU-14)
IN A
+IN A
V+
+IN B
IN B
OUT B
OUT D
IN D
+IN D
V
+IN C
IN C
OUT C
OUT A
AD8618
1
2
3
4
5
6
7
14
13
12
11
10
9
8
04648-0-049
Figure 4. 14-Lead SOIC (R-14)
GENERAL DESCRIPTION
The AD8616/AD8618 are dual/quad, rail-to-rail, input and
output, single-supply amplifiers featuring very low offset
voltage, wide signal bandwidth, and low input voltage and
current noise. The parts use a patented trimming technique that
achieves superior precision without laser trimming. The
AD8616/AD8618 are fully specified to operate from 2.7 V to
5 V single supplies.
The combination of 20 MHz bandwidth, low offset, low noise,
and very low input bias current make these amplifiers useful in
a wide variety of applications. Filters, integrators, photodiode
amplifiers, and high impedance sensors all benefit from the
combination of performance features. AC applications benefit
from the wide bandwidth and low distortion. The AD8616/
AD8618 offer the highest output drive capability of the
DigiTrim
TM
family, which is excellent for audio line drivers and
other low impedance applications.
Applications for the parts include portable and low powered
instrumentation, audio amplification for portable devices,
portable phone headsets, bar code scanners, and multipole
filters. The ability to swing rail to rail at both the input and
output enables designers to buffer CMOS ADCs, DACs, ASICs,
and other wide output swing devices in single-supply systems.
The AD8616/AD8618 are specified over the extended industrial
(40C to +125C) temperature range. The AD8616 is available
in 8-lead MSOP and narrow SOIC surface mount packages; the
MSOP version is available in tape and reel only. The AD8618 is
available in 14-lead SOIC and 14-lead TSSOP packages.
AD8616/AD8618
Rev. A | Page 2 of 16
TABLE OF CONTENTS
Specifications..................................................................................... 3
V
S
= 5 V.......................................................................................... 3
V
S
= 2.7 V....................................................................................... 4
Absolute Maximum Ratings............................................................ 5
Thermal Resistance ...................................................................... 5
ESD Caution.................................................................................. 5
Typical Performance Characteristics ............................................. 6
Applications..................................................................................... 12
Input Overvoltage Protection ................................................... 12
Output Phase Reversal............................................................... 12
Driving Capacitive Loads.......................................................... 12
Overload Recovery Time .......................................................... 13
D/A Conversion ......................................................................... 13
Low Noise Applications ............................................................. 13
High Speed Photodiode Preamplifier...................................... 14
Active Filters ............................................................................... 14
Power Dissipation ...................................................................... 14
Power Calculations for Varying or Unknown Loads............. 15
Outline Dimensions ....................................................................... 16
Ordering Guide .......................................................................... 16
REVISION HISTORY
4/04--Data Sheet Changed from Rev. 0 to Rev. A
Added AD8618................................................................Universal
Updated Outline Dimensions ................................................... 16
1/04--Revision 0: Initial Version
AD8616/AD8618
Rev. A | Page 3 of 16
SPECIFICATIONS
V
S
= 5 V
@V
CM
= V
S
/2, T
A
= 25C, unless otherwise noted.
Table 1.
Parameter
Symbol Conditions
Min Typ Max Unit
INPUT CHARACTERISTICS
Offset Voltage
V
OS
V
S
= 3.5 V @ V
CM
= 0.5 V and 3.0 V
23
65
V
V
CM
= 0 V to 5 V
80
500
V
-40C
<
T
A
< +125C
800
V
Offset Voltage Drift
V
OS
/T -40C
<
T
A
< +125C
1.5
7
V/C
Input Bias Current
I
B
0.2
1
pA
-40C
<
T
A
< +85C
50
pA
-40C
<
T
A
< +125C
500
pA
Input Offset Current
I
OS
0.1
0.5
pA
-40C
<
T
A
< +85C
50
pA
-40C
<
T
A
< +125C
250
pA
Input Voltage Range
0
5
V
Common-Mode Rejection Ratio
CMRR
V
CM
= 0 V to 4.5 V
80
100
dB
Large Signal Voltage Gain
A
VO
R
L
= 2 k, V
O
= 0.5 V to 5 V
105
1500
V/mV
Input Capacitance
C
DIFF
2.6
pF
C
CM
10
pF
OUTPUT CHARACTERISTICS
Output Voltage High
V
OH
I
L
= 1 mA
4.98
4.99
V
I
L
= 10 mA
4.88
4.92
V
-40C
<
T
A
< +125C
4.7
V
Output Voltage Low
V
OL
I
L
= 1 mA
7.5
15
mV
I
L
= 10 mA
70
100
mV
-40C
<
T
A
< +125C
200
mV
Output Current
I
OUT
150
mA
Closed-Loop Output Impedance
Z
OUT
f = 1 MHz, A
V
= 1
3
POWER SUPPLY
Power Supply Rejection Ratio
PSRR
V
S
= 2.7 V to 5.5 V
70
90
dB
Supply Current per Amplifier
I
SY
V
O
= 0 V
1.7
2.0
mA
-40C
<
T
A
< +125C
2.5
mA
DYNAMIC PERFORMANCE
Slew Rate
SR
R
L
= 2 k
12
V/s
Settling Time
t
s
To 0.01%
<0.5
s
Gain Bandwidth Product
GBP
24
MHz
Phase Margin
O
73
Degrees
NOISE PERFORMANCE
Peak-to-Peak Noise
e
n
p-p
0.1 Hz to 10 Hz
2.4
V
Voltage Noise Density
e
n
f = 1 kHz
8
nV/Hz
f = 10 kHz
6
nV/Hz
Current Noise Density
i
n
f = 1 kHz
0.05
pA/Hz
Channel Separation
Cs
f = 10 kHz
115
dB
f = 100 kHz
110
dB
AD8616/AD8618
Rev. A | Page 4 of 16
V
S
= 2.7 V
@V
CM
= V
S
/2, T
A
= 25C, unless otherwise noted.
Table 2.
Parameter
Symbol Conditions
Min Typ Max Unit
INPUT CHARACTERISTICS
Offset Voltage
V
OS
V
S
= 3.5 V @ V
CM
= 0.5 V and 3.0 V
23
65
V
V
CM
= 0 V to 2.7 V
80
500
V
-40C
<
T
A
< +125C
800
V
Offset Voltage Drift
V
OS
/T -40C
<
T
A
< +125C
1.5
7
V/C
Input Bias Current
I
B
0.2
1
pA
-40C
<
T
A
< +85C
50
pA
-40C
<
T
A
< +125C
500
pA
Input Offset Current
I
OS
0.1
0.5
pA
-40C
<
T
A
< +85C
50
pA
-40C
<
T
A
< +125C
250
pA
Input Voltage Range
0
2.7
V
Common-Mode Rejection Ratio
CMRR
V
CM
= 0 V to 2.7 V
84
100
dB
Large Signal Voltage Gain
A
VO
R
L
= 2 k, V
O
= 0.5 V to 2.2 V
55
150
V/mV
Input Capacitance
C
DIFF
2.6
pF
C
CM
10
pF
OUTPUT CHARACTERISTICS
Output Voltage High
V
OH
I
L
= 1 mA
2.65
2.68
V
-40C
<
T
A
< +125C
2.6
V
Output Voltage Low
V
OL
I
L
= 1 mA
11
25
mV
-40C
<
T
A
< +125C
30
mV
Output Current
I
OUT
50
mA
Closed-Loop Output Impedance
Z
OUT
f = 1 MHz, A
V
= 1
3
POWER SUPPLY
Power Supply Rejection Ratio
PSRR
V
S
= 2.7 V to 5.5 V
70
90
dB
Supply Current per Amplifier
I
SY
V
O
= 0 V
1.7
2
mA
-40C
<
T
A
< +125C
2.5
mA
DYNAMIC PERFORMANCE
Slew Rate
SR
R
L
= 2 k
12
V/s
Settling Time
t
s
To 0.01%
<0.3
s
Gain Bandwidth Product
GBP
22
MHz
Phase Margin
O
50
Degrees
NOISE PERFORMANCE
Peak-to-Peak Noise
e
n
p-p
0.1 Hz to 10 Hz
2.1
V
Voltage Noise Density
e
n
f = 1 kHz
8
nV/Hz
f = 10 kHz
6
nV/Hz
Current Noise Density
i
n
f = 1 kHz
0.05
pA/Hz
Channel Separation
Cs
f = 10 kHz
115
dB
f = 100 kHz
110
dB
AD8616/AD8618
Rev. A | Page 5 of 16
ABSOLUTE MAXIMUM RATINGS
Table 3. AD8616/AD8618 Stress Ratings
Parameter Rating
Supply Voltage
6 V
Input Voltage
GND to V
S
Differential Input Voltage
3 V
Ouput Short-Circuit Duration to GND
Indefinite
Storage Temperature
65C to +150C
Operating Temperature Range
40C to +125C
Lead Temperature Range (Soldering 60 sec)
300C
Junction Temperature
150C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
THERMAL RESISTANCE
JA
is specified for the worst-case conditions, i.e.,
JA
is specified
for device soldered in circuit board for surface-mount packages.
Table 4.
Package Type
JA
JC
Unit
8-Lead MSOP (RM)
210
45
C/W
8-Lead SOIC (R)
158
43
C/W
14-Lead SOIC (R)
120
36
C/W
14-Lead TSSOP (RU)
180
35
C/W
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
AD8616/AD8618
Rev. A | Page 6 of 16
TYPICAL PERFORMANCE CHARACTERISTICS
0
200
600
1400
1800
2200
1000
400
1200
1600
2000
800
NUMBE
R OF AMP
L
IFIE
RS
700
500
300
100
100
300
500
700
OFFSET VOLTAGE (
V)
04648-0-003
V
S
= 5V
T
A
= 25
C
V
CM
= 0V TO 5V
Figure 5. Input Offset Voltage Distribution
0
2
6
14
18
22
10
4
12
16
20
8
NUMBE
R O
F
AMP
L
IFIE
RS
0
2
4
6
8
10
12
TCV
OS
(
V/
C)
V
S
= 2.5V
T
A
= 40
C TO +125
C
V
CM
= 0V
04648-0-004
Figure 6. Offset Voltage Drift Distribution
400
500
300
200
100
0
100
200
300
400
500
IN
PU
T OFFSET VOLTA
GE (
V)
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
COMMON-MODE VOLTAGE (V)
V
S
= 5V
T
A
= 25
C
04648-0-005
Figure 7. Input Offset Voltage vs. Common-Mode Voltage
(200 Units, Five Wafer Lots Including Process Skews)
0
50
100
150
200
250
300
350
INP
U
T BIAS
CURRE
NT (
p
A)
0
25
50
75
100
125
TEMPERATURE (
C)
V
S
= 2.5V
04648-0-006
Figure 8. Input Bias Current vs. Temperature
V
SY
V
OUT
(mV
)
0.1
1
10
100
1000
0.001
0.01
0.1
1
10
100
LOAD CURRENT (mA)
V
S
= 5V
T
A
= 25
C
SINK
SOURCE
04648-0-007
Figure 9. Output Voltage to Supply Rail vs. Load Current
0
20
40
60
80
100
120
OUTPUT VOLTAGE (mV)
40 25 10
5
20
35
50
65
80
95
110 125
TEMPERATURE (
C)
04648-0-008
V
S
= 5V
1mA LOAD
10mA LOAD
Figure 10. Output Voltage Swing vs. Temperature
AD8616/AD8618
Rev. A | Page 7 of 16
40
20
0
20
40
60
80
100
G
A
IN (
d
B)
90
45
0
45
90
135
180
225
PH
A
SE (
D
egrees)
1k
10k
100k
1M
10M
100M
FREQUENCY (Hz)
V
S
= 2.5V
T
A
= 25
C
M
= 74
04648-0-009
Figure 11. Open-Loop Gain and Phase vs. Frequency
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
OUTPUT SW
ING (V p-p)
FREQUENCY (Hz)
10k
1k
100k
1M
10M
V
S
= 5.0V
V
IN
= 4.9V p-p
T
A
= 25
C
R
L
= 2k
A
V
= 1
04648-0-010
Figure 12. Closed-Loop Output Voltage Swing
0
10
20
30
40
50
60
70
80
90
100
OUTPUT IMPE
DANCE
(
)
1k
10k
100k
1M
10M
100M
FREQUENCY (Hz)
A
V
= 100
A
V
= 1
A
V
= 10
V
S
= 2.5V
04648-0-011
Figure 13. Output Impedance vs. Frequency
0
20
40
60
80
100
120
CMRR (
d
B)
FREQUENCY (Hz)
10k
1k
100k
1M
10M
V
S
= 2.5V
D8616-0-012
Figure 14. Common-Mode Rejection Ratio vs. Frequency
0
20
40
60
80
100
120
P
S
RR (dB)
FREQUENCY (Hz)
10k
1k
100k
1M
10M
V
S
= 2.5V
04648-0-013
Figure 15. PSRR vs. Frequency
5
0
10
15
20
25
30
35
40
45
50
SM
A
L
L SIGN
A
L
OVER
SH
OOT
(
%
)
CAPACITANCE (pF)
10
100
1000
04648-0-014
V
S
= 5V
R
L
=
T
A
= 25
C
A
V
= 1
+OS
OS
Figure 16. Small-Signal Overshoot vs. Load Capacitance
AD8616/AD8618
Rev. A | Page 8 of 16
0
0.4
0.8
0.6
0.2
1.2
1.0
S
U
P
P
L
Y
CURRE
NT P
E
R
AMP
L
IFIE
R (
m
A)
1.6
1.4
2.0
1.8
2.4
2.2
40 25 10
5
20
35
50
65
80
95
110 125
TEMPERATURE (
C)
V
S
= 2.7V
V
S
= 5V
04648-0-015
Figure 17. Supply Current vs. Temperature
200
0
400
600
800
1000
1200
1400
1600
1800
2000
S
U
P
P
L
Y
CURRE
NT P
E
R
AMP
L
IFIE
R (
A)
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
SUPPLY VOLTAGE (V)
04648-0-016
Figure 18. Supply Current vs. Supply Voltage
0
9
18
27
36
45
VOLTA
GE N
O
ISE D
E
N
S
ITY (
n
V/
H
z
)
54
63
72
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1.0
FREQUENCY (kHz)
V
S
= 5V
MKR @ 8.72
04648-0-017
Figure 19. Voltage Noise Density vs. Frequency
0
7
14
21
28
35
42
49
56
V
S
= 5V
MKR @ 6.70
04648-0-018
VOLTA
GE N
O
ISE D
E
N
S
ITY (
n
V/
H
z
)
0
1
2
3
4
5
6
7
8
9
10
FREQUENCY (kHz)
Figure 20. Voltage Noise Density vs. Frequency
VOLTA
GE (
50mV/D
I
V)
TIME (1
s/DIV)
V
S
= 5V
R
L
= 10k
C
L
= 200pF
A
V
= 1
04648-0-019
Figure 21. Small-Signal Transient Response
VO
LT
AG
E (
500mV/D
I
V)
TIME (1
s/DIV)
V
S
= 5V
R
L
= 10k
C
L
= 200pF
A
V
= 1
04648-0-020
Figure 22. Large-Signal Transient Response
AD8616/AD8618
Rev. A | Page 9 of 16
THD+
N (%)
0.0001
0.01
0.001
0.1
FREQUENCY (Hz)
20
100
1k
20k
04648-0-021
V
S
= 2.5V
V
IN
= 0.5V rms
A
V
= 1
BW = 22kHz
R
L
= 100k
Figure 23. THD + N
VOLTA
GE (
2
V/D
I
V)
V
S
= 2.5V
V
IN
= 2V p-p
A
V
= 10
TIME (200ns/DIV)
04648-0-022
Figure 24. Settling Time
VOLTAGE (1
V/D
I
V)
TIME (1s/DIV)
04648-0-023
V
S
= 2.7V
Figure 25. 0.1 Hz to 10 Hz Input Voltage Noise
0
200
400
600
800
1000
1200
1400
NUMBE
R O
F
AMP
L
IFIE
RS
700
500
300
100
100
300
500
700
OFFSET VOLTAGE (
V)
04648-0-024
V
S
= 2.7V
T
A
= 25
C
V
CM
= 0V TO 2.7V
Figure 26. Input Offset Voltage Distribution
400
500
300
200
100
0
100
200
300
400
500
IN
PU
T OFFSET VOLTA
GE (
V)
0
0.3
0.6
0.9
1.2
1.5
1.8
2.1
2.4
2.7
COMMON-MODE VOLTAGE (V)
V
S
= 2.7V
T
A
= 25
C
04648-0-025
Figure 27. Input Offset Voltage vs. Common-Mode Voltage
(200 Units, Five Wafer Lots Including Process Skews)
400
500
300
200
100
0
100
200
300
400
500
INPUT OFFSET VOLTAGE (
V)
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
COMMON-MODE VOLTAGE (V)
V
S
= 3.5V
T
A
= 25
C
04648-0-026
Figure 28. Input Offset Voltage vs. Common-Mode Voltage
(200 Units, Five Wafer Lots Including Process Skews)
AD8616/AD8618
Rev. A | Page 10 of 16
V
SY
V
OUT
(mV
)
0.1
1
10
100
1000
LOAD CURRENT (mA)
0.01
0.001
0.1
1
10
V
S
= 2.7V
T
A
= 25
C
SINK
SOURCE
04648-0-027
Figure 29. Output Voltage to Supply Rail vs. Load Current
0
2
4
6
8
10
12
14
16
18
OUTPUT VOLTAGE (mV)
40 25 10
5
20
35
50
65
80
95
110 125
TEMPERATURE (C)
V
S
= 2.7V
V
OH
@ 1mA LOAD
V
OL
@ 1mA LOAD
04648-0-028
Figure 30. Output Voltage Swing vs. Temperature
100
80
60
40
20
0
20
40
60
80
100
GAIN (
d
B)
225
180
135
90
45
0
45
90
135
180
225
PH
A
SE (
D
egrees)
V
S
=
1.35V
T
A
= 25
C
M
= 51
04648-0-029
FREQUENCY (Hz)
100k
1k
1M
10M
100M
10k
Figure 31. Open-Loop Gain and Phase vs. Frequency
0
0.3
0.6
0.9
1.2
1.5
1.8
2.1
2.4
2.7
OUTPUT SW
ING (V p-p)
FREQUENCY (Hz)
10k
1k
100k
1M
10M
V
S
= 2.7V
V
IN
= 2.6V p-p
T
A
= 25
C
R
L
= 2k
A
V
= 1
04648-0-030
Figure 32. Closed-Loop Output Voltage Swing vs. Frequency
5
0
10
15
20
25
30
35
40
45
50
SM
A
L
L SIGN
A
L
OVER
SH
OOT
(
%
)
CAPACITANCE (pF)
10
100
1000
V
S
= 1.35V
R
L
=
T
A
= 25
C
A
V
= 1
+OS
OS
04648-0-0331
Figure 33. Small-Signal Overshoot vs. Load Capacitance
0
8
16
24
32
40
VOLTA
GE N
O
ISE D
E
N
S
ITY (
n
V/ H
z
)
48
56
64
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1.0
FREQUENCY (kHz)
V
S
= 2.7V
MKR @ 7.47
04648-0-032
Figure 34. Voltage Noise Density vs. Frequency
AD8616/AD8618
Rev. A | Page 11 of 16
0
6
12
18
24
30
VOLTA
GE N
O
ISE D
E
N
S
ITY (
n
V/
H
z
)
36
42
48
0
1
2
3
4
5
6
7
8
9
10
FREQUENCY (kHz)
04648-0-033
V
S
= 2.7V
MKR @ 5.91
Figure 35. Voltage Noise Density vs. Frequency
VOLTA
GE (
50mV/D
I
V)
TIME (1
s/DIV)
V
S
= 2.7V
R
L
= 10k
C
L
= 200pF
A
V
= 1
04648-0-034
Figure 36. Small-Signal Transient Response
VO
LTA
G
E
(
500mV/D
I
V)
TIME (1
s/DIV)
V
S
= 2.7V
R
L
= 10k
C
L
= 200pF
A
V
= 1
04648-0-035
Figure 37. Large-Signal Transient Response
AD8616/AD8618
Rev. A | Page 12 of 16
APPLICATIONS
INPUT OVERVOLTAGE PROTECTION
The AD8616/AD8618 have internal protective circuitry that
allows voltages exceeding the supply to be applied at the input.
It is recommended, however, not to apply voltages that exceed
the supplies by more than 1.5 V at either input of the amplifier.
If a higher input voltage is applied, series resistors should be
used to limit the current flowing into the inputs.
The input current should be limited to <5 mA. The extremely
low input bias current allows the use of larger resistors, which
allows the user to apply higher voltages at the inputs. The use of
these resistors adds thermal noise, which contributes to the
overall output voltage noise of the amplifier.
For example, a 10 k resistor has less than 13 nV/Hz of
thermal noise and less than 10 nV of error voltage at room
temperature.
OUTPUT PHASE REVERSAL
The AD8616/AD8618 are immune to phase inversion, a
phenomenon that occurs when the voltage applied at the input
of the amplifier exceeds the maximum input common mode.
Phase reversal can cause permanent damage to the amplifier
and lock-ups to systems with feedback loops.
VOLTA
G
E (
2
V/D
I
V)
TIME (2ms/DIV)
04648-0-036
V
IN
V
OUT
V
S
= 2.5V
V
IN
= 6V p-p
A
V
= 1
R
L
= 10k
Figure 38. No Phase Reversal
DRIVING CAPACITIVE LOADS
Although the AD8616/AD8618 are capable of driving capacitive
loads of up to 500 pF without oscillating, a large amount of
overshoot is present when operating at frequencies above
100 kHz. This is especially true when the amplifier is configured
in positive unity gain (worst case). When such large capacitive
loads are required, the use of external compensation is highly
recommended. This reduces the overshoot and minimizes
ringing, which in turn improves the frequency response of the
AD8616/AD8618. One simple technique for compensation is
the snubber, which consists of a simple RC network. With this
circuit in place, output swing is maintained and the amplifier is
stable at all gains.
Figure 40 shows the implementation of the snubber, which
reduces overshoot by more than 30% and eliminates ringing,
which can cause instability. Using the snubber does not recover
the loss of bandwidth incurred from a heavy capacitive load.
VO
L
T
A
G
E
(
100mV/D
I
V)
TIME (2
s/DIV)
V
S
= 2.5V
A
V
= 1
C
L
= 500pF
04648-0-037
Figure 39. Driving Heavy Capacitive Loads without Compensation
V+
200
500pF
500pF
V
V
CC
V
EE
200mV
+
04648-0-038
+
Figure 40. Snubber Network
VO
L
T
A
G
E
(
100mV/D
I
V)
TIME (10
s/DIV)
V
S
= 2.5V
A
V
= 1
R
S
= 200
C
S
= 500pF
C
L
= 500pF
04648-0-039
Figure 41. Driving Heavy Capacitive Loads Using the Snubber Network
AD8616/AD8618
Rev. A | Page 13 of 16
OVERLOAD RECOVERY TIME
Overload recovery time is the time it takes the output of the
amplifier to come out of saturation and recover to its linear
region. Overload recovery is particularly important in
applications where small signals must be amplified in the
presence of large transients. Figure 42 and Figure 43 show the
positive and negative overload recovery times of the AD8616. In
both cases, the time elapsed before the AD8616 comes out of
saturation is less than 1 s. In addition, the symmetry between
the positive and negative recovery times allows for excellent
signal rectification without distortion to the output signal.
TIME (1
s/DIV)
V
S
= 2.5V
R
L
= 10k
A
V
= 100
V
IN
= 50mV
50mV
+2.5V
0V
0V
04648-0-040
Figure 42. Positive Overload Recovery
TIME (1
s/DIV)
V
S
= 2.5V
R
L
= 10k
A
V
= 100
V
IN
= 50mV
+50mV
2.5V
0V
0V
04648-0-041
Figure 43. Negative Overload Recovery
D/A CONVERSION
The AD8616 can be used at the output of high resolution DACs.
Their low offset voltage, fast slew rate, and fast settling time
make the parts suitable to buffer voltage output or current
output DACs.
Figure 44 shows an example of the AD8616 at the output of the
AD5542. The AD8616's rail-to-rail output and low distortion
help maintain the accuracy needed in data acquisition systems
and automated test equipment.
AD5542
V
OUT
UNIPOLAR
OUTPUT
AGND
DGND
REFS
1/2
AD8616
REFF
V
DD
SERIAL
INTERFACE
0.1
F
0.1
F
10
F
5V
2.5V
+
CS
DIN
SCLK
LDAC*
04648-
0-
042
Figure 44. Buffering DAC Output
LOW NOISE APPLICATIONS
Although the AD8618 typically has less than 8 nV/Hz of
voltage noise density at 1 kHz, it is possible to reduce it further.
A simple method is to connect the amplifiers in parallel, as
shown in Figure 45. The total noise at the output is divided by
the square root of the number of amplifiers. In this case, the
total noise is approximately 4 nV/Hz at room temperature.
The 100 resistor limits the current and provides an effective
output resistance of 50 .
V
R3
100
R1
10
V+
V
IN
04648-0-043
3
2
1
R2
1k
V
R6
100
R4
10
V+
3
2
1
R5
1k
V
R9
100
R7
10
V+
3
2
1
R8
1k
V
R12
100
R10
10
V+
3
2
1
R11
1k
V
OUT
Figure 45. Noise Reduction
AD8616/AD8618
Rev. A | Page 14 of 16
HIGH SPEED PHOTODIODE PREAMPLIFIER
The AD8616/AD8618 are excellent choices for I-to-V
conversions. The very low input bias, low current noise, and
high unity gain bandwidth of the parts make them suitable,
especially for high speed photodiode preamps.
In high speed photodiode applications, the diode is operated in
a photoconductive mode (reverse biased). This lowers the
junction capacitance at the expense of an increase in the
amount of dark current that flows out of the diode.
The total input capacitance, C1, is the sum of the diode
capacitance and that of the op amp. This creates a feedback pole
and causes degradation of the phase margin, making the op
amp unstable. It is therefore necessary to use a capacitor in the
feedback to compensate for this pole.
To get the maximum signal bandwidth, select
U
f
2
R
1
C
2
C
=
2
where f
U
is the unity gain bandwidth of the amplifier.
V
2.5V
V+
+2.5V
R2
C2
C
IN
C
D
R
SH
I
D
V
BIAS
04648-0-044
+
Figure 46. High Speed Photodiode Preamplifier
ACTIVE FILTERS
The low input bias current and high unity gain bandwidth of
the AD8616 make it an excellent choice for precision filter
design.
Figure 47 shows the implementation of a second-order low-pass
filter. The Butterworth response has a corner frequency of
100 kHz and a phase shift of 90. The frequency response is
shown in Figure 48.
V
V
EE
V+
V
CC
2nF
1nF
1.1k
1.1k
V
IN
04648-0-045
Figure 47. Second-Order Low-Pass Filter
40
30
20
10
0
10
GAIN (dB)
1
0.1
10
100
1k
10k
100k
1M
FREQUENCY (Hz)
04648-0-046
Figure 48. Second-Order Butterworth Low-Pass Filter Frequency Response
POWER DISSIPATION
Although the AD8616/AD8618 are capable of providing load
currents to 150 mA, the usable output load current drive
capability is limited to the maximum power dissipation allowed
by the device package used. In any application, the absolute
maximum junction temperature for the AD8616/AD8618 is
150C; this should never be exceeded because the device could
suffer premature failure. Accurately measuring power dissipa-
tion of an integrated circuit is not always a straightforward
exercise; Figure 49 has been provided as a design aid for setting
a safe output current drive level or selecting a heat sink for the
package options available on the AD8616.
POW
E
R
D
I
SSIPA
TION
(
W
)
TEMPERATURE (
C)
0
0
0.5
1.0
1.5
20
40
60
80
120
100
140
04648-0-047
SOIC
MSOP
Figure 49. Maximum Power Dissipation vs. Ambient Temperature
AD8616/AD8618
Rev. A | Page 15 of 16
These thermal resistance curves were determined using the
AD8616 thermal resistance data for each package and a
maximum junction temperature of 150C. The following
formula can be used to calculate the internal junction
temperature of the AD8616/AD8618 for any application:
T
J
= P
DISS
JA
+ T
A
where:
T
J
= junction temperature;
P
DISS
= power dissipation;
JA
= package thermal resistance, junction-to-case; and
T
A
= ambient temperature of the circuit.
To calculate the power dissipated by the AD8616/AD8618, use
the following equation:
P
DISS
= I
LOAD
(V
S
V
OUT
)
where:
I
LOAD
= output load current;
V
S
= supply voltage; and
V
OUT
= output voltage.
The quantity within the parentheses is the maximum voltage
developed across either output transistor.
POWER CALCULATIONS FOR VARYING OR
UNKNOWN LOADS
Often, calculating power dissipated by an integrated circuit to
determine if the device is being operated in a safe range is not
as simple as it might seem. In many cases, power cannot be
directly measured. This may be the result of irregular output
waveforms or varying loads; indirect methods of measuring
power are required.
There are two methods to calculate power dissipated by an
integrated circuit. The first can be done by measuring the
package temperature and the board temperature. The other is to
directly measure the circuit's supply current.
Calculating Power by Measuring Ambient and Case
Temperature
Given the two equations for calculating junction temperature:
T
J
= T
A
+ P
JA
where:
T
J
= junction temperature;
T
A
= ambient temperature.
JA
= the junction-to-ambient thermal resistance.
T
J
= T
C
+ P
JC
where T
C
is case temperature and
JA
and
JC
are given in the
data sheet.
The two equations can be solved for P (power):
T
A
+ P
JA
= T
C
+ P
JC
P = (T
A
T
C
)/(
JC
JA
)
Once power has been determined, it is necessary to go back and
calculate the junction temperature to assure that it has not been
exceeded.
The temperature measurements should be directly on the
package and on a spot on the board that is near the package but
not touching it. Measuring the package could be difficult. A very
small bimetallic junction glued to the package could be used; an
infrared sensing device could be used if the spot size is small
enough.
Calculating Power by Measuring Supply Current
Power can be calculated directly if the supply voltage and
current are known. However, supply current may have a dc
component with a pulse into a capacitive load. This could make
rms current very difficult to calculate. This can be overcome by
lifting the supply pin and inserting an rms current meter into
the circuit. For this to work, the user must be sure that all of the
current is being delivered by the supply pin being measured.
This is usually a good method in a single-supply system;
however, if the system uses dual supplies, both supplies may
need to be monitored.
AD8616/AD8618
Rev. A | Page 16 of 16
OUTLINE DIMENSIONS
0.80
0.60
0.40
8
0
4
8
5
4.90
BSC
PIN 1
0.65 BSC
3.00
BSC
SEATING
PLANE
0.15
0.00
0.38
0.22
1.10 MAX
3.00
BSC
COPLANARITY
0.10
0.23
0.08
COMPLIANT TO JEDEC STANDARDS MO-187AA
Figure 50. 8-Lead Micro Small Outline Package [MSOP]
(RM-8)
Dimensions shown in millimeters
0.25 (0.0098)
0.17 (0.0067)
1.27 (0.0500)
0.40 (0.0157)
0.50 (0.0196)
0.25 (0.0099)
45
8
0
1.75 (0.0688)
1.35 (0.0532)
SEATING
PLANE
0.25 (0.0098)
0.10 (0.0040)
4
1
8
5
5.00 (0.1968)
4.80 (0.1890)
4.00 (0.1574)
3.80 (0.1497)
1.27 (0.0500)
BSC
6.20 (0.2440)
5.80 (0.2284)
0.51 (0.0201)
0.31 (0.0122)
COPLANARITY
0.10
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
COMPLIANT TO JEDEC STANDARDS MS-012AA
Figure 51. 8-Lead Standard Small Outline Package [SOIC]
(R-8)
Dimensions shown in millimeters and (inches)
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
COPLANARITY
0.10
14
8
7
1
6.20 (0.2441)
5.80 (0.2283)
4.00 (0.1575)
3.80 (0.1496)
8.75 (0.3445)
8.55 (0.3366)
1.27 (0.0500)
BSC
SEATING
PLANE
0.25 (0.0098)
0.10 (0.0039)
0.51 (0.0201)
0.31 (0.0122)
1.75 (0.0689)
1.35 (0.0531)
8
0
0.50 (0.0197)
0.25 (0.0098)
1.27 (0.0500)
0.40 (0.0157)
0.25 (0.0098)
0.17 (0.0067)
COMPLIANT TO JEDEC STANDARDS MS-012AB
45
Figure 52. 14-Lead Standard Small Outline Package [SOIC]
(R-14)
Dimensions shown in millimeters and (inches)
4.50
4.40
4.30
14
8
7
1
6.40
BSC
PIN 1
5.10
5.00
4.90
0.65
BSC
SEATING
PLANE
0.15
0.05
0.30
0.19
1.20
MAX
1.05
1.00
0.80
0.20
0.09
8
0
0.75
0.60
0.45
COPLANARITY
0.10
COMPLIANT TO JEDEC STANDARDS MO-153AB-1
Figure 53. 14-Lead Thin Shrink Small Outline Package [TSSOP]
(RU-14)
Dimensions shown in millimeters
ORDERING GUIDE
Model Temperature
Range
Package
Description
Package Outline
Branding Code
AD8616ARM-R2
40C to +125C
8-Lead MSOP
RM-8
BLA
AD8616ARM-REEL
40C to +125C
8-Lead MSOP
RM-8
BLA
AD8616AR
40C to +125C
8-Lead SOIC
R-8
AD8616AR-REEL
40C to +125C
8-Lead SOIC
R-8
AD8616AR-REEL7
40C to +125C
8-Lead SOIC
R-8
AD8618AR
40C to +125C
14-Lead SOIC
R-14
AD8618AR-REEL
40C to +125C
14-Lead SOIC
R-14
AD8618AR-REEL7
40C to +125C
14-Lead SOIC
R-14
AD8618ARU
40C to +125C
14-Lead TSSOP
RU-14
AR8618ARU-REEL
40C to +125C
14-Lead TSSOP
RU-14
2004 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D0464804/04(A)
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