REV. 0

Information furnished by Analog Devices is believed to be accurate and
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otherwise under any patent or patent rights of Analog Devices.

ADP1111

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700

World Wide Web Site: http://www.analog.com

Fax: 617/326-8703

© Analog Devices, Inc., 1996

Micropower, Step-Up/Step-Down SW

Regulator; Adjustable and Fixed 3.3 V, 5 V, 12 V

FUNCTIONAL BLOCK DIAGRAMS

DRIVER

LIM

SW1

SW2

GND

SET

GAIN BLOCK/
ERROR AMP

COMPARATOR

1.25V

REFERENCE

OSCILLATOR

ADP1111

DRIVER

LIM

SW1

SW2

GND

SET

GAIN BLOCK/
ERROR AMP

COMPARATOR

SENSE

1.25V

REFERENCE

OSCILLATOR

ADP1111-5
ADP1111-12

R2 220k

FEATURES
Operates from 2 V to 30 V Input Voltage Range
72 kHz Frequency Operation
Utilizes Surface Mount Inductors
Very Few External Components Required
Operates in Step-Up/Step-Down or Inverting Mode
Low Battery Detector
User Adjustable Current Limit
Internal 1 A Power Switch
Fixed or Adjustable Output Voltage
8-Pin DIP or SO-8 Package

APPLICATIONS
3 V to 5 V, 5 V to 12 V Step-Up Converters
9 V to 5 V, 12 V to 5 V Step-Down Converters
Laptop and Palmtop Computers
Cellular Telephones
Flash Memory VPP Generators
Remote Controls
Peripherals and Add-On Cards
Battery Backup Supplies
Uninterruptible Supplies
Portable Instruments

GENERAL DESCRIPTION

The ADP1111 is part of a family of step-up/step-down switch-
ing regulators that operates from an input voltage supply of 2 V
to 12 V in step-up mode and up to 30 V in step-down mode.
The ADP1111 can be programmed to operate in step-up/step-
down or inverting applications with only 3 external components.

The fixed outputs are 3.3 V, 5 V and 12 V; and an adjustable
version is also available. The ADP1111 can deliver 100 mA at
5 V from a 3 V input in step-up mode, or it can deliver 200 mA
at 5 V from a 12 V input in step-down mode.

Maximum switch current can be programmed with a single
resistor, and an open collector gain block can be arranged in
multiple configuration for low battery detection, as a post linear
regulator, undervoltage lockout, or as an error amplifier.

If input voltages are lower than 2 V, see the ADP1110.

REV. 0

ADP1111SPECIFICATIONS

Parameter

Conditions

Min

Typ

Max

Units

QUIESCENT CURRENT

Switch Off

300

500

INPUT VOLTAGE

Step-Up Mode

2.0

12.6

Step-Down Mode

30.0

COMPARATOR TRIP POINT

VOLTAGE

ADP1111

1.20

1.25

1.30

OUTPUT SENSE VOLTAGE

ADP1111-3.3

OUT

3.13

3.30

3.47

ADP1111-5

4.75

5.00

5.25

ADP1111-12

11.40

12.00

12.60

COMPARATOR HYSTERESIS

ADP1111

12.5

OUTPUT HYSTERESIS

ADP1111-3.3

ADP1111-5

ADP1111-12

120

OSCILLATOR FREQUENCY

OSC

kHz

DUTY CYCLE

Full Load

SWITCH ON TIME

LIM

Tied to V

SW SATURATION VOLTAGE

= +25

STEP-UP MODE

= 3.0 V, I

= 650 mA

SAT

0.5

0.65

= 5.0 V, I

= 1 A

0.8

1.0

STEP-DOWN MODE

= 12 V, I

= 650 mA

1.1

1.5

FEEDBACK PIN BIAS CURRENT

ADP1111 V

= 0 V

160

300

SET PIN BIAS CURRENT

SET

= V

REF

SET

270

400

GAIN BLOCK OUTPUT LOW

SINK

= 300

SET

= 1.00 V

0.15

0.4

REFERENCE LINE REGULATION

5 V

30 V

0.02

0.075

%/V

2 V

5 V

0.4

%/V

GAIN BLOCK GAIN

= 100 k

1000

6000

V/V

CURRENT LIMIT

= +25

220

from I

LIM

to V

LIM

400

CURRENT LIMIT TEMPERATURE

COEFFICIENT

0.3

SWITCH OFF LEAKAGE CURRENT

= +25

Measured at SW1 Pin
V

SW1

= 12 V

MAXIMUM EXCURSION BELOW GND

= +25

SW1

A, Switch Off

400

350

NOTES

This specification guarantees that both the high and low trip points of the comparator fall within the 1.20 V to 1.30 V range.

The output voltage waveform will exhibit a sawtooth shape due to the comparator hysteresis. The output voltage on the fixed output versions will always be within

the specified range.

100 k

resistor connected between a 5 V source and the AO pin.

All limits at temperature extremes are guaranteed via correlation using standard statistical methods.

Specifications subject to change without notice.

(0 C

+70 C, V

= 3 V unless otherwise noted)

ADP1111

REV. 0

CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the ADP1111 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.

ORDERING GUIDE

Model

Output Voltage

Package*

ADP1111AN

ADJ

N-8

ADP1111AR

ADJ

SO-8

ADP1111AN-3.3

3.3 V

N-8

ADP1111AR-3.3

3.3 V

SO-8

ADP1111AN-5

5 V

N-8

ADP1111AR-5

5 V

SO-8

ADP1111AN-12

12 V

N-8

ADP1111AR-12

12 V

SO-8

*N = Plastic DIP, SO = Small Outline Package.

ABSOLUTE MAXIMUM RATINGS

Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36 V
SW1 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50 V
SW2 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . 0.5 V to V

Feedback Pin Voltage (ADP1111) . . . . . . . . . . . . . . . . . 5.5 V
Switch Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.5 A
Maximum Power Dissipation . . . . . . . . . . . . . . . . . . 500 mW
Operating Temperature Range
ADP1111A . . . . . . . . . . . . . . . . . . . . . . . . . . . 0

C to +70

Storage Temperature Range . . . . . . . . . . . . . 65

C to 150

Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . . . 300

PIN DESCRIPTIONS

Mnemonic

Function

LIM

For normal conditions this pin is connected to
V

. When lower current is required, a resistor

should be connected between I

LIM

and V

Limiting the switch current to 400 mA is achieved
by connecting a 220

resistor.

Input Voltage.

SW1

Collector Node of Power Transistor. For step-
down configuration, connect to V

. For step-up

configuration, connect to an inductor/diode.

SW2

Emitter Node of Power Transistor. For step-
down configuration, connect to inductor/diode.
For step-up configuration, connect to ground.
Do not allow this pin to go more than a diode
drop below ground.

GND

Ground.

Auxiliary Gain (GB) Output. The open collector
can sink 300

A. It can be left open if unused.

SET

Gain Amplifier Input. The amplifier's positive
input is connected to SET pin and its negative
input is connected to the 1.25 V reference. It can
be left open if unused.

FB/SENSE

On the ADP1111 (adjustable) version this pin
is connected to the comparator input. On the
ADP1111-3.3, ADP1111-5 and ADP1111-12,
the pin goes directly to the internal application
resistor that sets output voltage.

WARNING!

ESD SENSITIVE DEVICE

TYPICAL APPLICATION

10µF
(OPTIONAL)

LIM

SW1

SW2

GND

ADP1111AR-5

SUMIDA

CD54-220K

22µH

5V
100mA

33µF

SENSE

INPUT

MBRS120T3

Figure 1. 3 V to 5 V Step-Up Converter

PIN CONFIGURATIONS

8-Lead Plastic DIP

8-Lead SOIC

(N-8)

(SO-8)

LIM

SW1

GND

SW2

*FIXED VERSIONS

FB (SENSE)*

SET

ADP1111

TOP VIEW

(Not to Scale)

LIM

SW1

GND

SW2

*FIXED VERSIONS

FB (SENSE)*

SET

ADP1111

TOP VIEW

(Not to Scale)

SWITCH

CURRENT A

1.4

0.1

0.2

0.4

0.6

0.8

1.0

1.2

0.6

0.4

0.2

1.0

0.8

= 2V

= 3V

= 5V

SATURATION VOLTAGE V

Figure 2. Saturation Voltage vs. I

SWITCH

Current in

Step-Up Mode

SWITCH

CURRENT A

2.0

0.1

0.2

0.4

0.6

0.8

0.9

1.8

0.6

0.4

0.2

1.6

1.4

= 12V

ON VOLTAGE V

1.2

1.0

0.8

Figure 3. Switch ON Voltage vs. I

SWITCH

Current In

Step-Down Mode

INPUT VOLTAGE V

1.5

1400

1200

1000

800

600

400

200

QUIESCENT CURRENT

QUIESCENT CURRENT µA

Figure 4. Quiescent Current vs. Input Voltage

ADP1111Typical Characteristics

REV. 0

INPUT VOLTAGE V

OSCILLATOR FREQUENCY kHz

OSCILLATOR FREQUENCY

Figure 5. Oscillator Frequency vs. Input Voltage

LIM

1.9

1.7

0.1

1000

100

1.5

1.3

0.5

1.1

0.9

0.7

0.3

SWITCH CURRENT A

STEP-DOWN WITH
V

= 12V

STEP-UP WITH
2V < V

< 5V

Figure 6. Maximum Switch Current vs. R

LIM

OSCILLATOR FREQUENCY kHz

TEMPERATURE C

40

OSCILLATOR FREQUENCY

Figure 7. Oscillator Frequency vs. Temperature

ADP1111

REV. 0

TEMPERATURE C

7.5

6.6

40

7.3

7.2

6.8

6.7

7.4

ON TIME

ON TIME µs

7.1

6.9

7.0

Figure 8. Switch ON Time vs. Temperature

TEMPERATURE C

40

DUTY CYCLE

DUTY CYCLE %

Figure 9. Duty Cycle vs. Temperature

TEMPERATURE C

0.6

0.5

40

0.3

0.2

0.1

0.4

= 3V

= 0.65A

SATURATION VOLTAGE V

Figure 10. Saturation Voltage vs. Temperature in Step-Up
Mode

TEMPERATURE C

1.10

1.05

0.80

40

1.00

0.85

= 12V

= 0.65A

ON VOLTAGE V

0.95

0.90

Figure 11. Switch ON Voltage vs. Temperature in Step-
Down Mode

TEMPERATURE C

500

40

300

250

200

100

400

350

QUIESCENT CURRENT

QUIESCENT CURRENT µA

150

450

Figure 12. Quiescent Current vs. Temperature

TEMPERATURE C

250

40

150

100

200

BIAS CURRENT

BIAS CURRENT µA

Figure 13. Feedback Bias Current vs. Temperature

ADP1111

REV. 0

THEORY OF OPERATION

The ADP1111 is a flexible, low-power, switch-mode power
supply (SMPS) controller. The regulated output voltage can be
greater than the input voltage (boost or step-up mode) or less
than the input (buck or step-down mode). This device uses a
gated-oscillator technique to provide very high performance
with low quiescent current.

A functional block diagram of the ADP1111 is shown on
the first page of this data sheet. The internal 1.25 V reference is
connected to one input of the comparator, while the other input
is externally connected (via the FB pin) to a feedback network
connected to the regulated output. When the voltage at the FB
pin falls below 1.25 V, the 72 kHz oscillator turns on. A driver
amplifier provides base drive to the internal power switch, and
the switching action raises the output voltage. When the voltage
at the FB pin exceeds 1.25 V, the oscillator is shut off. While
the oscillator is off, the ADP1111 quiescent current is only
300

A. The comparator includes a small amount of hysteresis,

which ensures loop stability without requiring external compo-
nents for frequency compensation.

The maximum current in the internal power switch can be set
by connecting a resistor between V

and the I

LIM

pin. When the

maximum current is exceeded, the switch is turned OFF. The
current limit circuitry has a time delay of about 1

s. If an

external resistor is not used, connect I

LIM

to V

. Further

information on I

LIM

is included in the "APPLICATIONS"

section of this data sheet.

The ADP1111 internal oscillator provides 7

s ON and 7

OFF times that are ideal for applications where the ratio
between V

and V

OUT

is roughly a factor of two (such as

converting +3 V to + 5 V). However, wider range conversions
(such as generating +12 V from a +5 V supply) can easily be
accomplished.

An uncommitted gain block on the ADP1111 can be connected
as a low-battery detector. The inverting input of the gain block
is internally connected to the 1.25 V reference. The noninverting
input is available at the SET pin. A resistor divider, connected
between V

and GND with the junction connected to the SET

pin, causes the AO output to go LOW when the low battery set
point is exceeded. The AO output is an open collector NPN
transistor that can sink 300

The ADP1111 provides external connections for both the
collector and emitter of its internal power switch that permit
both step-up and step-down modes of operation. For the step-
up mode, the emitter (Pin SW2) is connected to GND, and the
collector (Pin SW1) drives the inductor. For step-down mode,
the emitter drives the inductor while the collector is connected
to V

The output voltage of the ADP1111 is set with two external
resistors. Three fixed-voltage models are also available:
ADP11113.3 (+3.3 V), ADP11115 (+5 V) and ADP111112
(+12 V). The fixed-voltage models are identical to the
ADP1111, except that laser-trimmed voltage-setting resistors
are included on the chip. On the fixed-voltage models of the
ADP1111, simply connect the feedback pin (Pin 8) directly to
the output voltage.

COMPONENT SELECTION
General Notes on Inductor Selection

When the ADP1111 internal power switch turns on, current
begins to flow in the inductor. Energy is stored in the inductor
core while the switch is on, and this stored energy is transferred
to the load when the switch turns off. Since both the collector
and the emitter of the switch transistor are accessible on the
ADP1111, the output voltage can be higher, lower, or of
opposite polarity than the input voltage.

To specify an inductor for the ADP1111, the proper values of
inductance, saturation current and dc resistance must be
determined. This process is not difficult, and specific equations
for each circuit configuration are provided in this data sheet. In
general terms, however, the inductance value must be low
enough to store the required amount of energy (when both
input voltage and switch ON time are at a minimum) but high
enough that the inductor will not saturate when both V

and

switch ON time are at their maximum values. The inductor
must also store enough energy to supply the load, without
saturating. Finally, the dc resistance of the inductor should be
low so that excessive power will not be wasted by heating the
windings. For most ADP1111 applications, an inductor of
15

H to 100

H with a saturation current rating of 300 mA to

1 A and dc resistance <0.4

is suitable. Ferrite-core inductors

that meet these specifications are available in small, surface-
mount packages.

To minimize Electro-Magnetic Interference (EMI), a toroid or
pot-core type inductor is recommended. Rod-core inductors are
a lower-cost alternative if EMI is not a problem.

CALCULATING THE INDUCTOR VALUE

Selecting the proper inductor value is a simple three step
process:

1. Define the operating parameters: minimum input voltage,

maximum input voltage, output voltage and output current.

2. Select the appropriate conversion topology (step-up, step-

down, or inverting).

3. Calculate the inductor value using the equations in the

following sections.

TEMPERATURE C

350

300

0
40

200

150

100

250

BIAS CURRENT

BIAS CURRENT µA

Figure 14. Set Pin Bias Current vs. Temperature

ADP1111

REV. 0

INDUCTOR SELECTIONSTEP-UP CONVERTER

In a step-up or boost converter (Figure 18), the inductor must
store enough power to make up the difference between the input
voltage and the output voltage. The power that must be stored
is calculated from the equation:

OUT

IN(MIN )

(

)

OUT

(

)

(Equation 1)

where V

is the diode forward voltage (0.5 V for a 1N5818

Schottky). Because energy is only stored in the inductor while
the ADP1111 switch is ON, the energy stored in the inductor
on each switching cycle must be equal to or greater than:

OSC

(Equation 2)

in order for the ADP1111 to regulate the output voltage.

When the internal power switch turns ON, current flow in the
inductor increases at the rate of:

( )

R't
L

(Equation 3)

where L is in Henrys and R' is the sum of the switch equivalent
resistance (typically 0.8

at +25

C) and the dc resistance of

the inductor. In most applications, the voltage drop across the
switch is small compared to V

so a simpler equation can be

used:

( )

(Equation 4)

Replacing `t' in the above equation with the ON time of the
ADP1111 (7

s, typical) will define the peak current for a given

inductor value and input voltage. At this point, the inductor
energy can be calculated as follows:

PEAK

(Equation 5)

As previously mentioned, E

must be greater than P

OSC

that the ADP1111 can deliver the necessary power to the load.
For best efficiency, peak current should be limited to 1 A or
less. Higher switch currents will reduce efficiency because of
increased saturation voltage in the switch. High peak current
also increases output ripple. As a general rule, keep peak current
as low as possible to minimize losses in the switch, inductor and
diode.

In practice, the inductor value is easily selected using the
equations above. For example, consider a supply that will
generate 12 V at 40 mA from a 9 V battery, assuming a 6 V
end-of-life voltage. The inductor power required is, from
Equation 1:

12V

0.5V

(

)

40 mA

(

)

260 mW

On each switching cycle, the inductor must supply:

OSC

260 mW

72 kHz

3.6

Since the required inductor power is fairly low in this example,
the peak current can also be low. Assuming a peak current of
500 mA as a starting point, Equation 4 can be rearranged to
recommend an inductor value:

L(MAX )

500 mA

Substituting a standard inductor value of 68

H with 0.2

resistance will produce a peak switch current of:

PEAK

1.0

587 mA

Once the peak current is known, the inductor energy can be
calculated from Equation 5:

(

)

587 mA

(

)

11.7

Since the inductor energy of 11.7

J is greater than the P

OSC

requirement of 3.6

J, the 68

H inductor will work in this

application. By substituting other inductor values into the same
equations, the optimum inductor value can be selected.

When selecting an inductor, the peak current must not exceed
the maximum switch current of 1.5 A. If the equations shown
above result in peak currents > 1.5 A, the ADP1110 should be
considered. Since this device has a 70% duty cycle, more energy
is stored in the inductor on each cycle. This results is greater
output power.

The peak current must be evaluated for both minimum and
maximum values of input voltage. If the switch current is high
when V

is at its minimum, the 1.5 A limit may be exceeded at

the maximum value of V

. In this case, the ADP1111's current

limit feature can be used to limit switch current. Simply select a
resistor (using Figure 6) that will limit the maximum switch
current to the I

PEAK

value calculated for the minimum value of

. This will improve efficiency by producing a constant I

PEAK

as V

increases. See the "Limiting the Switch Current" section

of this data sheet for more information.

Note that the switch current limit feature does not protect the
circuit if the output is shorted to ground. In this case, current is
only limited by the dc resistance of the inductor and the forward
voltage of the diode.

INDUCTOR SELECTIONSTEP-DOWN CONVERTER

The step-down mode of operation is shown in Figure 19.

Unlike the step-up mode, the ADP1111's power switch does not
saturate when operating in the step-down mode; therefore,
switch current should be limited to 650 mA in this mode. If the
input voltage will vary over a wide range, the I

LIM

pin can be

used to limit the maximum switch current. Higher switch
current is possible by adding an external switching transistor as
shown in Figure 21.

The first step in selecting the step-down inductor is to calculate
the peak switch current as follows:

PEAK

2 I

OUT

(Equation 6)

where DC = duty cycle (0.5 for the ADP1111)

= voltage drop across the switch

= diode drop (0.5 V for a 1N5818)

OUT

= output current

OUT

= the output voltage

= the minimum input voltage

ADP1111

REV. 0

As previously mentioned, the switch voltage is higher in step-
down mode than in step-up mode. V

is a function of switch

current and is therefore a function of V

, L, time and V

OUT

For most applications, a V

value of 1.5 V is recommended.

The inductor value can now be calculated:

IN MIN

(

)

OUT

PEAK

(Equation 7)

where t

= switch ON time (7

s).

If the input voltage will vary (such as an application that must
operate from a 9 V, 12 V or 15 V source), an R

LIM

resistor

should be selected from Figure 6. The R

LIM

resistor will keep

switch current constant as the input voltage rises. Note that
there are separate R

LIM

values for step-up and step-down modes

of operation.

For example, assume that +5 V at 300 mA is required from a
+12 V to +24 V source. Deriving the peak current from
Equation 6 yields:

PEAK

300 mA

0.5

1.5

0.5

600 mA

Then, the peak current can be inserted into Equation 7 to
calculate the inductor value:

1.5

600 mA

Since 64

H is not a standard value, the next lower standard

value of 56

H would be specified.

To avoid exceeding the maximum switch current when the
input voltage is at +24 V, an R

LIM

resistor should be specified.

Using the step-down curve of Figure 6, a value of 560

will

limit the switch current to 600 mA.

INDUCTOR SELECTIONPOSITIVE-TO-NEGATIVE
CONVERTER

The configuration for a positive-to-negative converter using the
ADP1111 is shown in Figure 22. As with the step-up converter,
all of the output power for the inverting circuit must be supplied
by the inductor. The required inductor power is derived from
the formula:

P =

OUT

(

)

(

)

(Equation 8)

The ADP1111 power switch does not saturate in positive-to-
negative mode. The voltage drop across the switch can be
modeled as a 0.75 V base-emitter diode in series with a 0.65

resistor. When the switch turns on, inductor current will rise at
a rate determined by:

( )

R't
L

(Equation 9)

where: R' = 0.65

+ R

L(DC)

= V

0.75 V

For example, assume that a 5 V output at 50 mA is to be
generated from a +4.5 V to +5.5 V source. The power in the
inductor is calculated from Equation 8:

5V|

0.5V|

(

)

50 mA

(

)

275 mW

During each switching cycle, the inductor must supply the
following energy:

OSC

275 mW

72 kHz

3.8

Using a standard inductor value of 56

H with 0.2

resistance will produce a peak switch current of:

PEAK

4.5V

0.75V

0.65

0.2

0.85

445 mA

Once the peak current is known, the inductor energy can be
calculated from (Equation 9):

(

)

445 mA

(

)

5.54

Since the inductor energy of 5.54

J is greater than the P

OSC

requirement of 3.82

J, the 56

H inductor will work in this

application.

The input voltage only varies between 4.5 V and 5.5 V in this
application. Therefore, the peak current will not change enough
to require an R

LIM

resistor and the I

LIM

pin can be connected

directly to V

. Care should be taken, of course, to ensure that

the peak current does not exceed 650 mA.

CAPACITOR SELECTION

For optimum performance, the ADP1111's output capacitor
must be selected carefully. Choosing an inappropriate capacitor
can result in low efficiency and/or high output ripple.

Ordinary aluminum electrolytic capacitors are inexpensive but
often have poor Equivalent Series Resistance (ESR) and
Equivalent Series Inductance (ESL). Low ESR aluminum
capacitors, specifically designed for switch mode converter
applications, are also available, and these are a better choice
than general purpose devices. Even better performance can be
achieved with tantalum capacitors, although their cost is higher.
Very low values of ESR can be achieved by using OS-CON
capacitors (Sanyo Corporation, San Diego, CA). These devices
are fairly small, available with tape-and-reel packaging and have
very low ESR.

The effects of capacitor selection on output ripple are demon-
strated in Figures 15, 16 and 17. These figures show the output
of the same ADP1111 converter that was evaluated with three
different output capacitors. In each case, the peak switch
current is 500 mA, and the capacitor value is 100

F. Figure 15

shows a Panasonic HF-series 16-volt radial cap. When the
switch turns off, the output voltage jumps by about 90 mV and
then decays as the inductor discharges into the capacitor. The
rise in voltage indicates an ESR of about 0.18

. In Figure 16,

the aluminum electrolytic has been replaced by a Sprague 293D
series, a 6 V tantalum device. In this case the output jumps
about 30 mV, which indicates an ESR of 0.06

. Figure 17

shows an OS-CON 16volt capacitor in the same circuit, and
ESR is only 0.02

ADP1111

REV. 0

Figure 15. Aluminum Electrolytic

Figure 16. Tantalum Electrolytic

Figure 17. OS-CON Capacitor

If low output ripple is important, the user should consider the
ADP3000. Because this device switches at 400 kHz, lower peak
current can be used. Also, the higher switching frequency
simplifies the design of the output filter. Consult the ADP3000
data sheet for additional details.

DIODE SELECTION

In specifying a diode, consideration must be given to speed,
forward voltage drop and reverse leakage current. When the
ADP1111 switch turns off, the diode must turn on rapidly if
high efficiency is to be maintained. Shottky rectifiers, as well as
fast signal diodes such as the 1N4148, are appropriate. The
forward voltage of the diode represents power that is not
delivered to the load, so V

must also be minimized. Again,

Schottky diodes are recommended. Leakage current is especially
important in low-current applications where the leakage can be
a significant percentage of the total quiescent current.

For most circuits, the 1N5818 is a suitable companion to the
ADP1111. This diode has a V

of 0.5 V at 1 A, 4

A to 10

leakage, and fast turn-on and turn-off times. A surface mount
version, the MBRS130T3, is also available.

For switch currents of 100 mA or less, a Shottky diode such as
the BAT85 provides a V

of 0.8 V at 100 mA and leakage less

than 1

A. A similar device, the BAT54, is available in a SOT23

package. Even lower leakage, in the 1 nA to 5 nA range, can be
obtained with a 1N4148 signal diode.

General purpose rectifiers, such as the 1N4001, are not suitable
for ADP1111 circuits. These devices, which have turn-on times
of 10

s or more, are far too slow for switching power supply

applications. Using such a diode "just to get started" will result
in wasted time and effort. Even if an ADP1111 circuit appears
to function with a 1N4001, the resulting performance will not
be indicative of the circuit performance when the correct diode
is used.

CIRCUIT OPERATION, STEP-UP (BOOST) MODE

In boost mode, the ADP1111 produces an output voltage that is
higher than the input voltage. For example, +12 V can be gener-
ated from a +5 V logic power supply or +5 V can be derived
from two alkaline cells (+3 V).

Figure 18 shows an ADP1111 configured for step-up operation.
The collector of the internal power switch is connected to the
output side of the inductor, while the emitter is connected to
GND. When the switch turns on, pin SW1 is pulled near
ground. This action forces a voltage across L1 equal to
V

CE(SAT)

, and current begins to flow through L1. This

current reaches a final value (ignoring second-order effects) of:

PEAK

CE (SAT )

where 7

s is the ADP1111 switch's "on" time.

LIM

SW1

GND

SW2

ADP1111

1N5818

OUT

(OPTIONAL)

Figure 18. Step-Up Mode Operation

When the switch turns off, the magnetic field collapses. The
polarity across the inductor changes, current begins to flow
through D1 into the load, and the output voltage is driven above
the input voltage.

The output voltage is fed back to the ADP1111 via resistors R1
and R2. When the voltage at pin FB falls below 1.25 V, SW1
turns "on" again, and the cycle repeats. The output voltage is
therefore set by the formula:

OUT

1.25 V

The circuit of Figure 18 shows a direct current path from V

OUT

, via the inductor and D1. Therefore, the boost converter

is not protected if the output is short circuited to ground.

ADP1111

REV. 0

CIRCUIT OPERATION, STEP DOWN (BUCK) MODE)

The ADP1111's step down mode is used to produce an output
voltage that is lower than the input voltage. For example, the
output of four NiCd cells (+4.8 V) can be converted to a +3 V
logic supply.

A typical configuration for step down operation of the ADP1111
is shown in Figure 19. In this case, the collector of the internal
power switch is connected to V

and the emitter drives the

inductor. When the switch turns on, SW2 is pulled up towards
V

. This forces a voltage across L1 equal to V

OUT

and causes current to flow in L1. This current reaches a final
value of:

PEAK

OUT

where 7

s is the ADP1111 switch's "on" time.

LIM

SW1

SW2 4

GND

SET

ADP1111

D1
1N5818

LIM

100

OUT

FB 8

Figure 19. Step-Down Mode Operation

When the switch turns off, the magnetic field collapses. The
polarity across the inductor changes, and the switch side of the
inductor is driven below ground. Schottky diode D1 then turns
on, and current flows into the load. Notice that the Absolute
Maximum Rating for the ADP1111's SW2 pin is 0.5 V below
ground. To avoid exceeding this limit, D1 must be a Schottky
diode. If a silicon diode is used for D1, Pin SW2 can go to
0.8 V, which will cause potentially damaging power dissipation
within the ADP1111.

The output voltage of the buck regulator is fed back to the
ADP1111's FB pin by resistors R1 and R2. When the voltage at
pin FB falls below 1.25 V, the internal power switch turns "on"
again, and the cycle repeats. The output voltage is set by the
formula:

OUT

1.25 V

When operating the ADP1111 in step-down mode, the output
voltage is impressed across the internal power switch's emitter-
base junction when the switch is off. To protect the switch, the
output voltage should be limited to 6.2 V or less. If a higher
output voltage is required, a Schottky diode should be placed in
series with SW2 as shown in Figure 20.

LIM

SW1

SW2

GND

ADP1111

OUT

D1, D2 = 1N5818 SCHOTTKY DIODES

Figure 20. Step-Down Model, V

OUT

> 6.2 V

If the input voltage to the ADP1111 varies over a wide range, a
current limiting resistor at Pin 1 may be required. If a particular
circuit requires high peak inductor current with minimum input
supply voltage, the peak current may exceed the switch maxi-
mum rating and/or saturate the inductor when the supply
voltage is at the maximum value. See the "Limiting the Switch
Current" section of this data sheet for specific recommendations.

INCREASING OUTPUT CURRENT IN THE STEP-DOWN
REGULATOR

Unlike the boost configuration, the ADP1111's internal power
switch is not saturated when operating in step-down mode. A
conservative value for the voltage across the switch in step-down
mode is 1.5 V. This results in high power dissipation within the
ADP1111 when high peak current is required. To increase the
output current, an external PNP switch can be added (Figure
21). In this circuit, the ADP1111 provides base drive to Q1
through R3, while R4 ensures that Q1 turns off rapidly. Because
the ADP1111's internal current limiting function will not work
in this circuit, R5 is provided for this purpose. With the value
shown, R5 limits current to 2 A. In addition to reducing power
dissipation on the ADP1111, this circuit also reduces the switch
voltage. When selecting an inductor value for the circuit of
Figure 21, the switch voltage can be calculated from the
formula:

= V

+ V

0.6 V + 0.4 V

1 V

Q1(SAT)

LIM

SW1

SW2

GND

SET

ADP1111

1N5821

0.3

INPUT

OUTPUT

330

220

MJE210

Figure 21. High Current Step-Down Operation

ADP1111

REV. 0

POSITIVE-TO-NEGATIVE CONVERSION

The ADP1111 can convert a positive input voltage to a negative
output voltage as shown in Figure 22. This circuit is essentially
identical to the step-down application of Figure 19, except that
the "output" side of the inductor is connected to power ground.
When the ADP1111's internal power switch turns off, current
flowing in the inductor forces the output (V

OUT

) to a negative

potential. The ADP1111 will continue to turn the switch on
until its FB pin is 1.25 V above its GND pin, so the output
voltage is determined by the formula:

OUT

1.25 V

LIM

SW1

SW2

GND

SET

ADP1111

1N5818

LIM

INPUT

OUTPUT

NEGATIVE
OUTPUT

Figure 22. Positive-to-Negative Converter

The design criteria for the step-down application also apply to
the positive-to-negative converter. The output voltage should be
limited to |6.2 V| unless a diode is inserted in series with the
SW2 pin (see Figure 20.) Also, D1 must again be a Schottky
diode to prevent excessive power dissipation in the ADP1111.

NEGATIVE-TO-POSITIVE CONVERSION

The circuit of Figure 23 converts a negative input voltage to a
positive output voltage. Operation of this circuit configuration is
similar to the step-up topology of Figure 18, except the current
through feedback resistor R2 is level-shifted below ground by a
PNP transistor. The voltage across R2 is V

OUT

BEQ1

. How-

ever, diode D2 level-shifts the base of Q1 about 0.6 V below
ground thereby cancelling the V

of Q1. The addition of D2

also reduces the circuit's output voltage sensitivity to tempera-
ture, which otherwise would be dominated by the 2 mV V

contribution of Q1. The output voltage for this circuit is
determined by the formula:

OUT

1.25 V

Unlike the positive step-up converter, the negative-to-positive
converter's output voltage can be either higher or lower than the
input voltage.

LIM

SW1

SW2

GND

SET

ADP1111

1N5818

10k

POSITIVE
OUTPUT

MJE210

LIM

NEGATIVE

INPUT

2N3906

Figure 23. ADP1111 Negative-to-Positive Converter

LIMITING THE SWITCH CURRENT

The ADP1111's R

LIM

pin permits the switch current to be

limited with a single resistor. This current limiting action occurs
on a pulse by pulse basis. This feature allows the input voltage
to vary over a wide range without saturating the inductor or
exceeding the maximum switch rating. For example, a particular
design may require peak switch current of 800 mA with a 2.0 V
input. If V

rises to 4 V, however, the switch current will

exceed 1.6 A. The ADP1111 limits switch current to 1.5 A and
thereby protects the switch, but the output ripple will increase.
Selecting the proper resistor will limit the switch current to
800 mA, even if V

increases. The relationship between R

LIM

and maximum switch current is shown in Figure 6.

The I

LIM

feature is also valuable for controlling inductor current

when the ADP1111 goes into continuous-conduction mode.

Table I. Component Selection for Typical Converters

Input

Output

Circuit

Inductor

Capacitor

Voltage

Current (mA)

Figure

Value

Part No.

Value

Notes

2 to 3.1

90 mA

CD75-150K

2 to 3.1

10 mA

CTX50-1

2 to 3.1

30 mA

CD75-150K

2 to 3.1

10 mA

CTX50-1

90 MA

CD75-330K

30 mA

CTX50-1

6.5 to 11

50 mA

12 to 20

300 mA

CTX50-4

20 to 30

300 mA

120

CTX100-4

7 mA

CTX50-4

250 mA

120

CTX100-4

100

NOTES
CD = Sumida.
CTX = Coiltronics.

*Add 47

from I

LIM

to V

**Add 220

from I

LIM

to V

ADP1111

REV. 0

This occurs in the step-up mode when the following condition is
met:

OUT

DIODE

where DC is the ADP1111's duty cycle. When this relationship
exists, the inductor current does not go all the way to zero
during the time that the switch is OFF. When the switch turns
on for the next cycle, the inductor current begins to ramp up
from the residual level. If the switch ON time remains constant,
the inductor current will increase to a high level (see Figure 24).
This increases output ripple and can require a larger inductor
and capacitor. By controlling switch current with the I

LIM

resistor, output ripple current can be maintained at the design
values. Figure 25 illustrates the action of the I

LIM

circuit.

Figure 24.

Figure 25.

The internal structure of the I

LIM

circuit is shown in Figure 26.

Q1 is the ADP1111's internal power switch that is paralleled by
sense transistor Q2. The relative sizes of Q1 and Q2 are scaled
so that I

is 0.5% of I

. Current flows to Q2 through an

internal 80

resistor and through the R

LIM

resistor. These two

resistors parallel the base-emitter junction of the oscillator-
disable transistor, Q3. When the voltage across R1 and R

LIM

exceeds 0.6 V, Q3 turns on and terminates the output pulse. If
only the 80

internal resistor is used (i.e. the I

LIM

pin is

connected directly to V

), the maximum switch current will be

1.5 A. Figure 6 gives R

LIM

values for lower current-limit values.

72kHz

OSC

POWER

SWITCH

SW2

SW1

LIM

DRIVER

(INTERNAL)

LIM

200

(EXTERNAL)

ADP1111

Figure 26. ADP1111 Current Limit Operation

The delay through the current limiting circuit is approximately
1

s. If the switch ON time is reduced to less than 3

s, accuracy

of the current trip-point is reduced. Attempting to program a
switch ON time of 1

s or less will produce spurious responses

in the switch ON time; however, the ADP1111 will still provide
a properly regulated output voltage.

PROGRAMMING THE GAIN BLOCK

The gain block of the ADP1111 can be used as a low-battery
detector, error amplifier or linear post regulator. The gain block
consists of an op amp with PNP inputs and an open-collector
NPN output. The inverting input is internally connected to the
ADP1111's 1.25 V reference, while the noninverting input is
available at the SET pin. The NPN output transistor will sink
about 300

Figure 27a shows the gain block configured as a low-battery
monitor. Resistors R1 and R2 should be set to high values to
reduce quiescent current, but not so high that bias current in
the SET input causes large errors. A value of 33 k

for R2 is a

good compromise. The value for R1 is then calculated from the
formula:

LOBATT

1.25 V

where V

LOBATT

is the desired low battery trip point. Since the

gain block output is an open-collector NPN, a pull-up resistor
should be connected to the positive logic power supply.

ADP1111

1.25V

REF

GND

47k

TO
PROCESSOR

BAT

SET

33k

R1= 

1.25V

35.1µA

= BATTERY TRIP POINT

Figure 27a. Setting the Low Battery Detector Trip Point

200mA/div

ADP1111

REV. 0

The circuit of Figure 27b may produce multiple pulses when
approaching the trip point due to noise coupled into the SET
input. To prevent multiple interrupts to the digital logic,
hysteresis can be added to the circuit (Figure 27). Resistor
RHYS, with a value of 1 M

to 10 M

, provides the hysteresis.

The addition of RHYS will change the trip point slightly, so the
new value for R1 will be:

LOBATT

1.25 V

HYS

where V

is the logic power supply voltage, R

is the pull-up

resistor, and R

HYS

creates the hysteresis.

ADP1111

1.25V

REF

GND

47k

TO
PROCESSOR

BAT

SET

HYS

33k

1.6M

Figure 27b.

APPLICATION CIRCUITS
All Surface Mount 3 V to 5 V Step-Up Converter

This is the most basic application (along with the basic step-
down configuration to follow) of the ADP1111. It takes full
advantage of surface mount packaging for all the devices used in
the design. The circuit can provide +5 V at 100 mA of output
current and can be operated off of battery power for use in
portable equipment.

33µF

OUTPUT

R3*

(OPTIONAL)

MBRS120T3

20µH

CTX20-4

(5V @ 100mA)

INPUT +3V

LIM

SW2

SW1

SENSE

GND

SET

ADP1111-5

Figure 28. All Surface Mount +3 V to +5 V Step-Up Converter

9 V to 5 V Step-Down Converter

This circuit uses a 9 V battery to generate a +5 V output. The
circuit will work down to 6.5 V, supplying 50 mA at this lower
limit. Switch current is limited to around 500 mA by the 100

resistor.

LIM

SW1

SW2

SENSE

GND

SET

ADP1111-5

22µF

D1
1N5818

LIM

100

15µH

CTX15-4

INPUT

OUTPUT
(9V

TO 5V @ 150mA,

6.5V

TO 5V @ 50mA)

Figure 29. 9 V to 5 V Step-Down Converter

20 V to 5 V Step-Down Converter

This circuit is similar to Figure 29, except it supplies much
higher output current and operates over a much wider range of
input voltage. As in the previous examples, switch current is
limited to 500 mA.

LIM

SW1

SW2

SENSE

GND

SET

ADP1111-5

47µF

D1
1N5818

LIM

100

68µH

CTX68-4

12V TO 28V

INPUT

OUTPUT
(+5V @ 300mA)

Figure 30. 20 V to 5 V Step-Down Converter

+5 V to 5 V Converter

This circuit is essentially identical to Figure 22, except it uses a
fixed-output version of the ADP1111 to simplify the design
somewhat.

LIM

SW1

SW2

GND

SET

ADP1111-5

33µF

D1
1N5818

LIM

100

33µH

CTX33-2

12V TO 28V

INPUT

5V
@ 75mA

SENSE

Figure 31. +5 V to 5 V Converter

ADP1111

REV. 0

Voltage-Controlled Positive-to-Negative Converter

By including an op amp in the feedback path, a simple positive-
to-negative converter can be made to give an output that is a
linear multiple of a controlling voltage, Vc. The op amp, an
OP196, rail-to-rail input and output amplifier, sums the
currents from the output and controlling voltage and drives the
FB pin either high or low, thereby controlling the on-board
oscillator. The 0.22

resistor limits the short-circuit current to

about 3 A and, along with the BAT54 Schottky diode, helps
limit the peak switch current over varying input voltages. The
external power switch features an active pull-up to speed up the
turn-off time of the switch. Although an IRF9530 was used in
the evaluation, almost any device that can handle at least 3 A of
peak current at a VDS of at least 50 V is suitable for use in this
application, provided that adequate attention is paid to power
dissipation. The circuit can deliver 2 W of output power with a
+6-volt input from a control voltage range of 0 V to 5 V.

LIM

SW1

SW2

GND

SET

ADP1111

47µF

35V

OUTPUT

LIM

INPUT

+5V TO +12V

0.22

200k

39k

1N4148

IRF9530

20µH

IN5821

OUT

= 5.13 *V

2W MAXIMUM OUTPUT

1N5231

CTX20-4

(0V TO +5V)

2N3904

BAT54

Figure 32. Voltage Controlled Positive-to-Negative
Converter

+3 V to 22 V LCD Bias Generator

This circuit uses an adjustable-output version of the ADP1111
to generate a +22.5 V reference output that is level-shifted to
give an output of 22 V. If operation from a +5 volt supply is
desired, change R1 to 47 ohms. The circuit will deliver 7 mA
with a 3 volt supply and 40 mA with a 5 volt supply.

0.1µF

OUTPUT

25µH

1N4148

1N5818

732k

42.2k

22µF

4.7

µF

+3V

2xAA

CELLS

LIM

100

22V OUTPUT
7mA @ 2V INPUT

L1 = CTX25-4

LIM

SW1

SW2

GND

SET

ADP1111

Figure 33. 3 V to 22 V LCD Bias Generator

High Power, Low Quiescent Current Step-Down Converter

By making use of the fact that the feedback pin directly controls
the internal oscillator, this circuit achieves a shutdown-like state
by forcing the feedback pin above the 1.25 V comparator
threshold. The logic level at the 1N4148 diode anode needs to
be at least 2 V for reliable standby operation.

The external switch driver circuit features an active pull-up
device, a 2N3904 transistor, to ensure that the power MOSFET
turns off quickly. Almost any power MOSFET will do as the
switch as long as the device can withstand the 18 volt V

and is

reasonably robust. The 0.22

resistor limits the short-circuit

current to about 3 A and, along with the BAT54 Schottky
diode, helps to limit the peak switch current over varying input
voltages.

220µF

IRF9540

D1
IN5821

LIM

INPUT

+8V TO +18V

0.22

1N4148

BAT54

2N3904

121k

40.2k

1N4148

+5V

500mA

OPERATE

/STANDBY

LIM

SW1

SW2

GND

SET

ADP1111

20µH

LI = COILTRONICS CTX20-4

Figure 34. High Power, Low Quiescent Current Step-Down
Converter

NOTES
1. All inductors referenced are Coiltronics CTX-series except

where noted.

2. If the source of power is more than an inch or so from the

converter, the input to the converter should be bypassed with
approximately 10

F of capacitance. This capacitor should

be a good quality tantalum or aluminum electrolytic.

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