FN9055.5

ISL6526

Single Synchronous Buck Pulse-Width

Modulation (PWM) Controller

The ISL6526 makes simple work out of implementing a
complete control and protection scheme for a DC-DC
stepdown converter. Designed to drive N-Channel
MOSFETs in a synchronous buck topology, the ISL6526
integrates the control, output adjustment, monitoring and
protection functions into a single package.

The ISL6526 provides simple, single feedback loop, voltage-
mode control with fast transient response. The output
voltage can be precisely regulated to as low as 0.8V, with a
maximum tolerance of ±1.5% over temperature and line
voltage variations. A fixed frequency oscillator reduces
design complexity, while balancing typical application cost
and efficiency.

The error amplifier features a 15MHz gain-bandwidth
product and 6V/

µs slew rate which enables high converter

bandwidth for fast transient performance. The resulting
PWM duty cycles range from 0% to 100%.

Protection from overcurrent conditions is provided by
monitoring the r

DS(ON)

of the upper MOSFET to inhibit PWM

operation appropriately. This approach simplifies the
implementation and improves efficiency by eliminating the
need for a current sense resistor.

Features

· Operates from 3.3V to 5V Input

· 0.8V to V

Output Range

- 0.8V Internal Reference
- ±1.5% Over Load, Line Voltage and Temperature

· Drives N-Channel MOSFETs

· Simple Single-Loop Control Design

- Voltage-Mode PWM Control

· Fast Transient Response

- High-Bandwidth Error Amplifier
- Full 0% to 100% Duty Cycle

· Lossless, Programmable Overcurrent Protection

- Uses Upper MOSFET's r

DS(on)

· Converter can Source and Sink Current

· Small Converter Size

- Internal Fixed Frequency Oscillator

- ISL6526: 300kHz
- ISL6526A: 600kHz

· Internal Soft-Start

· 14 Lead SOIC or 16 Lead, 5x5 QFN

· QFN Package:

- Compliant to JEDEC PUB95 MO-220 QFN - Quad Flat

No Leads - Package Outline

- Near Chip Scale Package footprint, which improves

PCB efficiency and has a thinner profile

· Pb-free Available

Applications

· Power Supplies for Microprocessors

- PCs
- Embedded Controllers

· Subsystem Power Supplies

- PCI/AGP/GTL+ Buses
- ACPI Power Control
- DDR SDRAM Bus Termination Supply

· Cable Modems, Set Top Boxes, and DSL Modems

· DSP and Core Communications Processor Supplies

· Memory Supplies

· Personal Computer Peripherals

· Industrial Power Supplies

· 3.3V-Input DC-DC Regulators

· Low-Voltage Distributed Power Supplies

Data Sheet

August 2004

CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.

1-888-INTERSIL or 321-724-7143

Intersil (and design) is a registered trademark of Intersil Americas Inc.

All other trademarks mentioned are the property of their respective owners.

Ordering Information

PART #

TEMP RANGE (°C)

PACKAGE

PKG DWG. #

ISL6526CBZ (Note 1)

0 to 70

14 Ld SOIC (Pb-free)

M14.15

ISL6526ACBZ (Note 1)

0 to 70

14 Ld SOIC (Pb-free)

M14.15

ISL6526CR

0 to 70

16 Ld 5x5 QFN

L16.5x5B

ISL6526ACR

0 to 70

16 Ld 5x5 QFN

L16.5x5B

ISL6526IB

-40 to 85

14 Ld SOIC

M14.15

ISL6526IBZ (Note 1)

-40 to 85

14 Ld SOIC (Pb-free)

M14.15

ISL6526AIB

-40 to 85

14 Ld SOIC

M14.15

ISL6526IRZ (Note 1)

-40 to 85

16 Ld 5x5 QFN (Pb-free) L16.5x5B

ISL6526AIR

-40 to 85

16 Ld 5x5 QFN

L16.5x5B

ISL6526AIRZ (Note 1)

-40 to 85

16 Ld 5x5 QFN (Pb-free) L16.5x5B

ISL6526EVAL1

ISL6526 SOIC Evaluation Board

ISL6526EVAL2

ISL6526 QFN Evaluation Board

ISL6526AEVAL1

ISL6526A SOIC Evaluation Board

ISL6526AEVAL2

ISL6526A QFN Evaluation Board

NOTES:

1. Intersil Pb-free products employ special Pb-free material sets; molding

compounds/die attach materials and 100% matte tin plate termination finish, which
is compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free
products are MSL classified at Pb-free peak reflow temperatures that meet or
exceed the Pb-free requirements of IPC/JEDEC J Std-020B.

2. Add "-T" suffix for Tape and Reel.

Pinouts

ISL6526

14 LEAD (SOIC)

TOP VIEW

ISL6526

16 LEAD 5X5 (QFN)

TOP VIEW

LGATE

CPVOUT

UGATE

PHASE

VCC

BOOT

GND

CT1

CT2

OCSET

CPGND

ENABLE

COMP

CPVOUT

CT1

CT2

OCSET

GND

UGA

BOOT

PHASE

VCC

CPGND

COMP

ENABLE

ISL6526

Absolute Maximum Ratings

Thermal Information

Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+7V
Absolute Boot Voltage, V

BOOT

. . . . . . . . . . . . . . . . . . . . . . . +15.0V

Upper Driver Supply Voltage, V

BOOT

- V

PHASE

. . . . . . . . . . . +6.0V

Input, Output or I/O Voltage . . . . . . . . . . . GND -0.3V to VCC +0.3V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2

Operating Conditions

Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . +3.3V ±10%
Ambient Temperature Range . . . . . . . . . . . . . . . . . . . .-40°C to 85°C
Junction Temperature Range. . . . . . . . . . . . . . . . . . .-40°C to 125°C

Thermal Resistance

(°C/W)

SOIC Package (Note 3) . . . . . . . . . . . .

N/A

QFN Package (Note 4). . . . . . . . . . . . .

Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . 150°C
Maximum Storage Temperature Range . . . . . . . . . . . -65°C to 150°C
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300°C

(SOIC - Lead Tips Only)

For Recommended soldering conditions see Tech Brief TB389.

CAUTION: Stresses above those listed in "Absolute Maximum Ratings" may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.

NOTES:

is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.

is measured in free air with the component mounted on a high effective thermal conductivity test board with "direct attach" features.

JC,

the

"case temp" is measured at the center of the exposed metal pad on the package underside. See Tech Brief TB379.

Electrical Specifications

Recommended Operating Conditions, unless otherwise noted V

= 3.3V±5% and T

= 25°C

PARAMETER

SYMBOL

TEST CONDITIONS

MIN

TYP

MAX

UNITS

VCC SUPPLY CURRENT

Nominal Supply

BIAS

6.1

6.9

7.7

POWER-ON RESET

Rising CPVOUT POR Threshold

POR

Commercial

4.25

4.30

4.42

Industrial

4.10

4.30

4.50

CPVOUT POR Threshold Hysteresis

0.3

0.6

0.9

OSCILLATOR

Frequency

OSC

IC = ISL6526C, Commercial

275

300

325

kHz

IC = ISL6526I, Industrial

250

300

340

kHz

IC = ISL6526AC, Commercial

575

600

625

kHz

IC = ISL6526AI, Industrial

550

600

640

kHz

Ramp Amplitude

OSC

1.5

P-P

REFERENCE

Reference Voltage Tolerance

1.5

Nominal Reference Voltage

REF

0.800

Charge Pump

Nominal Charge Pump Output

CPVOUT

VCC

= 3.3V, No Load

5.1

Charge Pump Output Regulation

ERROR AMPLIFIER

DC Gain

Guaranteed by Design

Gain-Bandwidth Product

GBWP

MHz

Slew Rate

µs

SOFT-START

Soft-Start Slew Rate

Commercial

6.2

7.3

Industrial

6.2

7.6

ISL6526

Functional Pin Description

14 LEAD (SOIC)

TOP VIEW

16 LEAD 5X5 (QFN)

TOP VIEW

VCC

This pin provides the bias supply for the ISL6526. Connect a
well-decoupled 3.3V supply to this pin.

COMP and FB

COMP and FB are the available external pins of the error
amplifier. The FB pin is the inverting input of the internal
error amplifier and the COMP pin is the error amplifier
output. These pins are used to compensate the voltage-
control feedback loop of the converter.

GND

This pin represents the signal and power ground for the IC.
Tie this pin to the ground island/plane through the lowest
impedance connection available.

PHASE

Connect this pin to the upper MOSFET's source. This pin is
used to monitor the voltage drop across the upper MOSFET
for overcurrent protection.

UGATE

Connect this pin to the upper MOSFET's gate. This pin
provides the PWM-controlled gate drive for the upper
MOSFET. This pin is also monitored by the adaptive shoot-
through protection circuitry to determine when the upper
MOSFET has turned off.

BOOT

This pin provides ground referenced bias voltage to the
upper MOSFET driver. A bootstrap circuit is used to create a
voltage suitable to drive a logic-level N-Channel MOSFET.

LGATE

Connect this pin to the lower MOSFET's gate. This pin
provides the PWM-controlled gate drive for the lower
MOSFET. This pin is also monitored by the adaptive shoot-
through protection circuitry to determine when the lower
MOSFET has turned off.

OCSET

Connect a resistor (R

OCSET

) from this pin to the drain of the

upper MOSFET (V

). R

OCSET

, an internal 20

µA current

source (I

OCSET

), and the upper MOSFET on-resistance

DS(ON)

) set the converter overcurrent (OC) trip point

according to the following equation:

An overcurrent trip cycles the soft-start function.

GATE DRIVERS

Upper Gate Source Current

UGATE-SRC

BOOT

- V

PHASE

= 5V, V

UGATE

= 4V

-1

Upper Gate Sink Current

UGATE-SNK

Lower Gate Source Current

LGATE-SRC

VCC

= 3.3V, V

LGATE

= 4V

-1

Lower Gate Sink Current

LGATE-SNK

PROTECTION/DISABLE

OCSET Current Source

OCSET

Commercial

µA

Industrial

µA

Disable Threshold

DISABLE

0.8

Electrical Specifications

Recommended Operating Conditions, unless otherwise noted V

= 3.3V±5% and T

= 25°C (Continued)

PARAMETER

SYMBOL

TEST CONDITIONS

MIN

TYP

MAX

UNITS

LGATE

CPVOUT

UGATE

PHASE

VCC

BOOT

GND

CT1

CT2

OCSET

CPGND

ENABLE

COMP

CPVOUT

CT1

CT2

OCSET

GND

PHASE

VCC

CPGND

ENABLE

PEAK

OCSET

DS ON

(

)

-------------------------------------------------

ISL6526

ENABLE

This pin is the open-collector enable pin. Pulling this pin to a
level below 0.8V will disable the controller. Disabling the
ISL6526 causes the oscillator to stop, the LGATE and UGATE
outputs to be held low, and the soft-start circuitry to re-arm.

CT1 and CT2

These pins are the connections for the external charge
pump capacitor. A minimum of a 0.1

µF ceramic capacitor is

recommended for proper operation of the IC.

CPVOUT

This pin represents the output of the charge pump. The
voltage at this pin is the bias voltage for the IC. Connect a
decoupling capacitor from this pin to ground. The value of
the decoupling capacitor should be at least 10x the value of
the charge pump capacitor. This pin may be tied to the
bootstrap circuit as the source for creating the BOOT
voltage.

CPGND

This pin represents the signal and power ground for the
charge pump. Tie this pin to the ground island/plane through
the lowest impedance connection available.

Functional Description

Initialization

The ISL6526 automatically initializes upon receipt of power.
Special sequencing of the input supplies is not necessary.
The Power-On Reset (POR) function continually monitors
the output voltage of the charge pump. During POR, the
charge pump operates on a free running oscillator. Once the
POR level is reached, the charge pump oscillator is synched
to the PWM oscillator. The POR function also initiates the
soft-start operation after the charge pump output voltage
exceeds its POR threshold.

Soft-Start

The POR function initiates the digital soft-start sequence.
The PWM error amplifier reference is clamped to a level
proportional to the soft-start voltage. As the soft-start voltage
slews up, the PWM comparator generates PHASE pulses of
increasing width that charge the output capacitor(s). This
method provides a rapid and controlled output voltage rise.
The soft-start sequence typically takes about 6.5ms.

Figure 1 shows the soft-start sequence for a typical application.
At t0, the +3.3V VCC voltage starts to ramp. At time t1, the
Charge Pump begins operation and the +5V CPVOUT IC bias
voltage starts to ramp up. Once the voltage on CPVOUT
crosses the POR threshold at time t2, the output begins the
soft-start sequence. The triangle waveform from the PWM
oscillator is compared to the rising error amplifier output
voltage. As the error amplifier voltage increases, the pulse-
width on the UGATE pin increases to reach the steady-state
duty cycle at time t3.

Shoot-Through Protection

A shoot-through condition occurs when both the upper
MOSFET and lower MOSFET are turned on simultaneously,
effectively shorting the input voltage to ground. To protect
the regulator from a shoot-through condition, the ISL6526
incorporates specialized circuitry which insures that the
complementary MOSFETs are not ON simultaneously.

The adaptive shoot-through protection utilized by the
ISL6526 looks at the lower gate drive pin, LGATE, and the
upper gate drive pin, UGATE, to determine whether a
MOSFET is ON or OFF. If the voltage from UGATE or from
LGATE to GND is less than 0.8V, then the respective
MOSFET is defined as being OFF and the complementary
MOSFET is turned ON. This method of shoot-through
protection allows the regulator to sink or source current.

Since the voltage of the lower MOSFET gate and the upper
MOSFET gate are being measured to determine the state of
the MOSFET, the designer is encouraged to consider the
repercussions of introducing external components between
the gate drivers and their respective MOSFET gates before
actually implementing such measures. Doing so may
interfere with the shoot-through protection.

Output Voltage Selection

The output voltage can be programmed to any level between
V

and the internal reference, 0.8V. An external resistor

divider is used to scale the output voltage relative to the
reference voltage and feed it back to the inverting input of
the error amplifier, see Figure 2. However, since the value of
R1 affects the values of the rest of the compensation
components, it is advisable to keep its value less than 5k

R4 can be calculated based on the following equation:

If the output voltage desired is 0.8V, simply route the output
back to the FB pin through R1, but do not populate R4.

FIGURE 1. SOFT-START INTERVAL

TIME

CPVOUT (5V)

VCC (3.3V)

OUT

(2.50V)

(1V/DIV)

R1 0.8V

OUT1

0.8V

--------------------------------------

ISL6526

Overcurrent Protection

The overcurrent function protects the converter from a shorted
output by using the upper MOSFET on-resistance, r

DS(ON)

, to

monitor the current. This method enhances the converter's
efficiency and reduces cost by eliminating a current sensing
resistor.

The overcurrent function cycles the soft-start function in a
hiccup mode to provide fault protection. A resistor (R

OCSET

)

programs the overcurrent trip level (see Typical Application
diagrams on pages 2 and 3). An internal 20

µA (typical) current

sink develops a voltage across R

OCSET

that is referenced to

. When the voltage across the upper MOSFET (also

referenced to V

) exceeds the voltage across R

OCSET

, the

overcurrent function initiates a soft-start sequence.

Figure 3 illustrates the protection feature responding to an
overcurrent event. At time t0, an overcurrent condition is
sensed across the upper MOSFET. As a result, the regulator
is quickly shutdown and the internal soft-start function begins
producing soft-start ramps. The delay interval seen by the
output is equivalent to three soft-start cycles. The fourth
internal soft-start cycle initiates a normal soft-start ramp of
the output, at time t1. The output is brought back into
regulation by time t2, as long as the overcurrent event has
cleared.

Had the cause of the overcurrent still been present after the
delay interval, the overcurrent condition would be sensed
and the regulator would be shut down again for another
delay interval of three soft-start cycles. The resulting hiccup
mode style of protection would continue to repeat
indefinitely.

The overcurrent function will trip at a peak inductor current
(I

PEAK)

determined by:

where I

OCSET

is the internal OCSET current source (20

µA

typical). The OC trip point varies mainly due to the MOSFET
r

DS(ON)

variations. To avoid overcurrent tripping in the

normal operating load range, find the R

OCSET

resistor from

the equation above with:

1. The maximum r

DS(ON)

at the highest junction

temperature.

2. The minimum I

OCSET

from the specification table.

3. Determine I

PEAK

for

where

I is the output inductor ripple current.

For an equation for the ripple current see the section under
component guidelines titled `Output Inductor Selection'.

A small ceramic capacitor should be placed in parallel with
R

OCSET

to smooth the voltage across R

OCSET

in the

presence of switching noise on the input voltage.

Current Sinking

The ISL6526 incorporates a MOSFET shoot-through
protection method which allows a converter to sink current
as well as source current. Care should be exercised when
designing a converter with the ISL6526 when it is known that
the converter may sink current.

When the converter is sinking current, it is behaving as a
boost converter that is regulating its input voltage. This
means that the converter is boosting current into the input
rail of the regulator. If there is nowhere for this current to go,

FIGURE 2. OUTPUT VOLTAGE SELECTION

OUT

+3.3V

OUT

ISL6526

UGATE

VCC

BOOT

COMP

PHASE

LGATE

CPVOUT

VIN

FIGURE 3. OVERCURRENT PROTECTION RESPONSE

TIME

OUT

(2.5V)

INTERNAL SOFT-START FUNCTION

DELAY INTERVAL

PEAK

OCSET

x R

OCSET

DS ON

(

)

-----------------------------------------------------

PEAK

OUT MAX

(

)

( )

----------

ISL6526

such as to other distributed loads on the rail or through a
voltage limiting protection device, the capacitance on this rail
will absorb the current. This situation will allow the voltage
level of the input rail to increase. If the voltage level of the rail
is boosted to a level that exceeds the maximum voltage
rating of any components attached to the input rail, then
those components may experience an irreversible failure or
experience stress that may shorten their lifespan. Ensuring
that there is a path for the current to flow other than the
capacitance on the rail will prevent this failure mode.

Application Guidelines

Layout Considerations

Layout is very important in high frequency switching
converter design. With power devices switching efficiently at
300kHz or 600kHz, the resulting current transitions from one
device to another causing voltage spikes across the
interconnecting impedances and parasitic circuit elements.
These voltage spikes can degrade efficiency, radiate noise
into the circuit, and lead to device overvoltage stress.
Careful component layout and printed circuit board design
minimizes the voltage spikes in the converters.

As an example, consider the turn-off transition of the PWM
MOSFET. Prior to turn-off, the MOSFET is carrying the full load
current. During turn-off, current stops flowing in the MOSFET
and is picked up by the lower MOSFET. Any parasitic
inductance in the switched current path generates a large
voltage spike during the switching interval. Careful component
selection, tight layout of the critical components, and short, wide
traces minimizes the magnitude of voltage spikes.

There are two sets of critical components in a DC-DC
converter using the ISL6526. The switching components are
the most critical because they switch large amounts of
energy, and therefore tend to generate large amounts of
noise. Next are the small signal components which connect
to sensitive nodes or supply critical bypass current and
signal coupling.

A multi-layer printed circuit board is recommended. Figure 4
shows the connections of the critical components in the
converter. Note that capacitors C

and C

OUT

could each

represent numerous physical capacitors. Dedicate one solid
layer, usually a middle layer of the PC board, for a ground
plane and make all critical component ground connections
with vias to this layer. Dedicate another solid layer as a
power plane and break this plane into smaller islands of
common voltage levels. Keep the metal runs from the
PHASE terminals to the output inductor short. The power
plane should support the input power and output power
nodes. Use copper filled polygons on the top and bottom
circuit layers for the phase nodes. Use the remaining printed
circuit layers for small signal wiring. The wiring traces from
the GATE pins to the MOSFET gates should be kept short
and wide enough to easily handle the 1A of drive current.

The switching components should be placed close to the
ISL6526 first. Minimize the length of the connections between
the input capacitors, C

, and the power switches by placing

them nearby. Position both the ceramic and bulk input
capacitors as close to the upper MOSFET drain as possible.
Position the output inductor and output capacitors between the
upper MOSFET and lower MOSFET and the load.

The critical small signal components include any bypass
capacitors, feedback components, and compensation
components. Position the bypass capacitor, C

, close to

the VCC pin with a via directly to the ground plane. Place the
PWM converter compensation components close to the FB
and COMP pins. The feedback resistors for both regulators
should also be located as close as possible to the relevant
FB pin with vias tied straight to the ground plane as required.

Feedback Compensation

Figure 5 highlights the voltage-mode control loop for a
synchronous-rectified buck converter. The output voltage
(V

OUT

) is regulated to the Reference voltage level. The

error amplifier (Error Amp) output (V

E/A

) is compared with

the oscillator (OSC) triangular wave to provide a pulse-

OUT

ISLAND ON POWER PLANE LAYER

ISLAND ON CIRCUIT PLANE LAYER

OUT

+3.3V V

KEY

COMP

ISL6526

UGATE

GND

CPVOUT

FIGURE 4. PRINTED CIRCUIT BOARD POWER PLANES

AND ISLANDS

BOOT

VIA CONNECTION TO GROUND PLANE

BOOT

PHASE

LGATE

PHASE

VCC

ISL6526

width modulated (PWM) wave with an amplitude of V

the PHASE node. The PWM wave is smoothed by the output
filter (L

and C

The modulator transfer function is the small-signal transfer
function of V

OUT

E/A

. This function is dominated by a DC

Gain and the output filter (L

and C

), with a double pole

break frequency at F

and a zero at F

ESR

. The DC Gain of

the modulator is simply the input voltage (V

) divided by the

peak-to-peak oscillator voltage

OSC

Modulator Break Frequency Equations

The compensation network consists of the error amplifier
(internal to the ISL6526) and the impedance networks Z

and Z

. The goal of the compensation network is to provide

a closed loop transfer function with the highest 0dB crossing
frequency (f

0dB

) and adequate phase margin. Phase margin

is the difference between the closed loop phase at f

0dB

and

180 degrees. The equations below relate the compensation
network's poles, zeros and gain to the components (R

, R

, C

, and C

) in Figure 5. Use these guidelines for

locating the poles and zeros of the compensation network:

1. Pick gain (R

) for desired converter bandwidth.

2. Place first zero below filter's double pole (~75% F

3. Place second zero at filter's double pole.
4. Place first pole at the ESR zero.
5. Place second pole at half the switching frequency.
6. Check gain against error amplifier's open-loop gain.
7. Estimate phase margin - repeat if necessary.

Compensation Break Frequency Equations

Figure 6 shows an asymptotic plot of the DC-DC converter's
gain vs frequency. The actual Modulator Gain has a high gain
peak due to the high Q factor of the output filter and is not
shown in Figure 6. Using the above guidelines should give a
Compensation Gain similar to the curve plotted. The open
loop error amplifier gain bounds the compensation gain.
Check the compensation gain at F

with the capabilities of

the error amplifier. The Closed Loop Gain is constructed on
the graph of Figure 6 by adding the Modulator Gain (in dB) to
the Compensation Gain (in dB). This is equivalent to
multiplying the modulator transfer function to the
compensation transfer function and plotting the gain.

The compensation gain uses external impedance networks
Z

and Z

to provide a stable, high bandwidth (BW) overall

loop. A stable control loop has a gain crossing with
-20dB/decade slope and a phase margin greater than 45
degrees. Include worst case component variations when
determining phase margin.

FIGURE 5. VOLTAGE-MODE BUCK CONVERTER

COMPENSATION DESIGN

OUT

REFERENCE

ESR

OSC

ERROR

AMP

PWM

DRIVER

(PARASITIC)

REFERENCE

COMP

OUT

ISL6526

COMPARATOR

DRIVER

DETAILED COMPENSATION COMPONENTS

PHASE

E/A

OSC

x L

x C

------------------------------------------

ESR

x ESR x C

-------------------------------------------

x R

(

) x C

-------------------------------------------------------

x R

x C

----------------------

---------------------------------------------------------

x R

x C

------------------------------------

----------------------------------

FIGURE 6. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN

100

-20

-40

-60

10M

100K

10K

100

OPEN LOOP

ERROR AMP GAIN

ESR

COMPENSATION

IN (

FREQUENCY (Hz)

GAIN

MODULATOR

GAIN

LOOP GAIN

VIN

VOSC

------------------

log

R2
R1

--------

log

ISL6526

Component Selection Guidelines

Charge Pump Capacitor Selection

A capacitor across pins CT1 and CT2 is required to create
the proper bias voltage for the ISL6526 when operating the
IC from 3.3V. Selecting the proper capacitance value is
important so that the bias current draw and the current
required by the MOSFET gates do not overburden the
capacitor. A conservative approach is presented in the
following equation.

Output Capacitor Selection

An output capacitor is required to filter the output and supply
the load transient current. The filtering requirements are a
function of the switching frequency and the ripple current.
The load transient requirements are a function of the slew
rate (di/dt) and the magnitude of the transient load current.
These requirements are generally met with a mix of
capacitors and careful layout.

Modern digital ICs can produce high transient load slew
rates. High frequency capacitors initially supply the transient
and slow the current load rate seen by the bulk capacitors.
The bulk filter capacitor values are generally determined by
the ESR (Effective Series Resistance) and voltage rating
requirements rather than actual capacitance requirements.

High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements.

Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The
bulk capacitor's ESR will determine the output ripple voltage
and the initial voltage drop after a high slew-rate transient. An
aluminum electrolytic capacitor's ESR value is related to the
case size with lower ESR available in larger case sizes.
However, the Equivalent Series Inductance (ESL) of these
capacitors increases with case size and can reduce the
usefulness of the capacitor to high slew-rate transient loading.
Unfortunately, ESL is not a specified parameter. Work with
your capacitor supplier and measure the capacitor's
impedance with frequency to select a suitable component. In
most cases, multiple electrolytic capacitors of small case size
perform better than a single large case capacitor.

Output Inductor Selection

The output inductor is selected to meet the output voltage
ripple requirements and minimize the converter's response
time to the load transient. The inductor value determines the
converter's ripple current and the ripple voltage is a function

of the ripple current. The ripple voltage and current are
approximated by the following equations:

Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values reduce
the converter's response time to a load transient.

One of the parameters limiting the converter's response to
a load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
ISL6526 will provide either 0% or 100% duty cycle in
response to a load transient. The response time is the time
required to slew the inductor current from an initial current
value to the transient current level. During this interval the
difference between the inductor current and the transient
current level must be supplied by the output capacitor.
Minimizing the response time can minimize the output
capacitance required.

The response time to a transient is different for the
application of load and the removal of load. The following
equations give the approximate response time interval for
application and removal of a transient load:

where: I

TRAN

is the transient load current step, t

RISE

is the

response time to the application of load, and t

FALL

is the

response time to the removal of load. The worst case
response time can be either at the application or removal of
load. Be sure to check both of these equations at the
minimum and maximum output levels for the worst case
response time.

Input Capacitor Selection

Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use small ceramic
capacitors for high frequency decoupling and bulk capacitors
to supply the current needed each time Q

turns on. Place the

small ceramic capacitors physically close to the MOSFETs
and between the drain of Q

and the source of Q

The important parameters for the bulk input capacitor are the
voltage rating and the RMS current rating. For reliable
operation, select the bulk capacitor with voltage and current
ratings above the maximum input voltage and largest RMS
current required by the circuit. The capacitor voltage rating
should be at least 1.25 times greater than the maximum
input voltage and a voltage rating of 1.5 times is a
conservative guideline. The RMS current rating requirement
for the input capacitor of a buck regulator is approximately
1/2 the DC load current.

PUMP

BiasAndGate

------------------------------------ 1.5

I =

- V

OUT

x L

OUT

I x ESR

RISE

L x I

TRAN

- V

OUT

FALL

L x I

TRAN

OUT

ISL6526

The maximum RMS current required by the regulator may be
closely approximated through the following equation:

For a through hole design, several electrolytic capacitors may
be needed. For surface mount designs, solid tantalum
capacitors can be used, but caution must be exercised with
regard to the capacitor surge current rating. These capacitors
must be capable of handling the surge-current at power-up.
Some capacitor series available from reputable manufacturers
are surge current tested.

MOSFET Selection/Considerations

The ISL6526 requires two N-Channel power MOSFETs.
These should be selected based upon r

DS(ON)

, gate supply

requirements, and thermal management requirements.

In high-current applications, the MOSFET power dissipation,
package selection and heatsink are the dominant design
factors. The power dissipation includes two loss components;
conduction loss and switching loss. The conduction losses are
the largest component of power dissipation for both the upper
and the lower MOSFETs. These losses are distributed between
the two MOSFETs according to duty factor. The switching
losses seen when sourcing current will be different from the
switching losses seen when sinking current. When sourcing
current, the upper MOSFET realizes most of the switching
losses. The lower switch realizes most of the switching losses
when the converter is sinking current (see equations on next
page). These equations assume linear voltage-current
transitions and do not adequately model power loss due the
reverse-recovery of the upper and lower MOSFET's body
diode. The gate-charge losses are dissipated by the ISL6526
and don't heat the MOSFETs. However, large gate-charge
increases the switching interval, t

which increases the

MOSFET switching losses. Ensure that both MOSFETs are
within their maximum junction temperature at high ambient
temperature by calculating the temperature rise according to
package thermal-resistance specifications. A separate heatsink
may be necessary depending upon MOSFET power, package
type, ambient temperature and air flow.

Given the reduced available gate bias voltage (5V), logic-
level or sub-logic-level transistors should be used for both
N-MOSFETs. Caution should be exercised with devices
exhibiting very low V

GS(ON)

characteristics. The shoot-

through protection present aboard the ISL6526 may be
circumvented by these MOSFETs if they have large parasitic
impedances and/or capacitances that would inhibit the gate
of the MOSFET from being discharged below its threshold
level before the complementary MOSFET is turned on.

Bootstrap Component Selection

External bootstrap components, a diode and capacitor, are
required to provide sufficient gate enhancement to the upper
MOSFET. The internal MOSFET gate driver is supplied by
the external bootstrap circuitry as shown in Figure 7. The
boot capacitor, C

BOOT

, develops a floating supply voltage

referenced to the PHASE pin. This supply is refreshed each
cycle, when D

BOOT

conducts, to a voltage of CPVOUT less

the boot diode drop, V

, plus the voltage rise across

LOWER

Just after the PWM switching cycle begins and the charge
transfer from the bootstrap capacitor to the gate capacitance
is complete, the voltage on the bootstrap capacitor is at its
lowest point during the switching cycle. The charge lost on
the bootstrap capacitor will be equal to the charge
transferred to the equivalent gate-source capacitance of the
upper MOSFET as shown:

where Q

GATE

is the maximum total gate charge of the upper

MOSFET, C

BOOT

is the bootstrap capacitance, V

BOOT1

the bootstrap voltage immediately before turn-on, and
V

BOOT2

is the bootstrap voltage immediately after turn-on.

The bootstrap capacitor begins its refresh cycle when the gate
drive begins to turn-off the upper MOSFET. A refresh cycle
ends when the upper MOSFET is turned on again, which
varies depending on the switching frequency and duty cycle.

RMSMAX

OUT

--------------

OUTMAX

------

OUT

L f

-----------------------------

OUT

--------------

LOWER

= Io

x r

DS(ON)

x (1 - D)

Where: D is the duty cycle = V

OUT

/ V

is the combined switch ON and OFF time, and

is the switching frequency.

Losses while Sourcing current

Losses while Sinking current

LOWER

DS ON

(

)

1 D

(

)

1
2

--- Io

UPPER

DS ON

(

)

1
2

--- Io

UPPER

= Io

x r

DS(ON)

x D

ISL6526

GND

LGATE

UGATE

PHASE

BOOT

NOTE:

NOTE:
V

G-S

= V

BOOT

UPPER

LOWER

FIGURE 7. UPPER GATE DRIVE BOOTSTRAP

G-S

= V

-V

CPVOUT

GATE

BOOT

BOOT1

BOOT2

(

)

ISL6526

The minimum bootstrap capacitance can be calculated by
rearranging the previous equation and solving for C

BOOT

Typical gate charge values for MOSFETs considered in
these types of applications range from 20 to 100nC. Since
the voltage drop across Q

LOWER

is negligible, V

BOOT1

simply V

CPVOUT

- V

. A schottky diode is recommended to

minimize the voltage drop across the bootstrap capacitor
during the on-time of the upper MOSFET. Initial calculations
with V

BOOT2

no less than 4V will quickly help narrow the

bootstrap capacitor range.

For example, consider an upper MOSFET is chosen with a
maximum gate charge, Q

, of 100nC. Limiting the voltage

drop across the bootstrap capacitor to 1V results in a value

of no less than 0.1

µF. The tolerance of the ceramic capacitor

should also be considered when selecting the final bootstrap
capacitance value.

A fast recovery diode is recommended when selecting a
bootstrap diode to reduce the impact of reverse recovery
charge loss. Otherwise, the recovery charge, Q

, would

have to be added to the gate charge of the MOSFET and
taken into consideration when calculating the minimum
bootstrap capacitance.

ISL6526 DC-DC Converter Application
Circuit

Figure 8 shows an application circuit of a DC-DC Converter.
Detailed information on the circuit, including a complete Bill-
of-Materials and circuit board description, can be found in
Application Note AN9994.

BOOT

GATE

BOOT1

BOOT2

-----------------------------------------------------

Component Selection Notes:

3,8,9

- Each 150

µF, Panasonic EEF-UE0J151R or Equivalent.

D1 - 30mA Schottky Diode, MA732 or Equivalent
L

- 1

µH Inductor, Panasonic P/N ETQ-P6F1ROSFA or Equivalent.

- Fairchild MOSFET; ITF86110DK8.

2.5V @ 5A

COMP

UGATE

PHASE

BOOT

GND

LGATE

ISL6526

8,9

OCSET

CPVOUT

CT1

CT2

CPGND

VCC

3.3V

GND

ENABLE

9.76k

6.49k

2.26k

124

0.1

µF

1000pF

µF

0.22

µF

0.1

µF

33pF

5600pF

8200pF

1.07k

CERAMIC

ENABLE

FIGURE 8. 3.3V TO 2.5V 5A DC-DC CONVERTER

ISL6526

ISL6526

Small Outline Plastic Packages (SOIC)

NOTES:

1. Symbols are defined in the "MO Series Symbol List" in Section 2.2 of

Publication Number 95.

2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension "D" does not include mold flash, protrusions or gate burrs.

Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006
inch) per side.

4. Dimension "E" does not include interlead flash or protrusions. Interlead

flash and protrusions shall not exceed 0.25mm (0.010 inch) per side.

5. The chamfer on the body is optional. If it is not present, a visual index

feature must be located within the crosshatched area.

6. "L" is the length of terminal for soldering to a substrate.
7. "N" is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. The lead width "B", as measured 0.36mm (0.014 inch) or greater

above the seating plane, shall not exceed a maximum value of
0.61mm (0.024 inch).

10. Controlling dimension: MILLIMETER. Converted inch dimensions

are not necessarily exact.

INDEX

AREA

-B-

0.25(0.010)

C A

B S

-A-

-C-

SEATING PLANE

0.10(0.004)

h x 45

0.25(0.010)

M14.15

(JEDEC MS-012-AB ISSUE C)

14 LEAD NARROW BODY SMALL OUTLINE PLASTIC
PACKAGE

SYMBOL

INCHES

MILLIMETERS

NOTES

MIN

MAX

MIN

MAX

0.0532

0.0688

1.35

1.75

0.0040

0.0098

0.10

0.25

0.013

0.020

0.33

0.51

0.0075

0.0098

0.19

0.25

0.3367

0.3444

8.55

8.75

0.1497

0.1574

3.80

4.00

0.050 BSC

1.27 BSC

0.2284

0.2440

5.80

6.20

0.0099

0.0196

0.25

0.50

0.016

0.050

0.40

1.27

Rev. 0 12/93

All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.

Intersil Corporation's quality certifications can be viewed at www.intersil.com/design/quality

Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.

For information regarding Intersil Corporation and its products, see www.intersil.com

ISL6526

Quad Flat No-Lead Plastic Package (QFN)
Micro Lead Frame Plastic Package (MLFP)

L16.5x5B

16 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
(COMPLIANT TO JEDEC MO-220VHHB ISSUE C)

SYMBOL

MILLIMETERS

NOTES

MIN

NOMINAL

MAX

0.80

0.90

1.00

0.05

1.00

0.20 REF

0.28

0.33

0.40

5, 8

5.00 BSC

4.75 BSC

2.95

3.10

3.25

7, 8

5.00 BSC

4.75 BSC

2.95

3.10

3.25

7, 8

0.80 BSC

0.25

0.35

0.60

0.75

L1 -

0.15

0.60

Rev. 1 10/02

NOTES:

1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
2. N is the number of terminals.
3. Nd and Ne refer to the number of terminals on each D and E.
4. All dimensions are in millimeters. Angles are in degrees.
5. Dimension b applies to the metallized terminal and is measured

between 0.15mm and 0.30mm from the terminal tip.

6. The configuration of the pin #1 identifier is optional, but must be

located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.

7. Dimensions D2 and E2 are for the exposed pads which provide

improved electrical and thermal performance.

8. Nominal dimensions are provided to assist with PCB Land Pattern

Design efforts, see Intersil Technical Brief TB389.

9. Features and dimensions A2, A3, D1, E1, P &

are present when

Anvil singulation method is used and not present for saw
singulation.

10. Depending on the method of lead termination at the edge of the

package, a maximum 0.15mm pull back (L1) maybe present. L
minus L1 to be equal to or greater than 0.3mm.

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