IRU3037 / IRU3037A

Rev. 2.8
03/10/03

www.irf.com

TYPICAL APPLICATION

DESCRIPTION

The IRU3037 controller IC is designed to provide a low
cost synchronous Buck regulator for on-board DC to DC
converter applications. With the migration of today's ASIC
products requiring low supply voltages such as 1.8V and
lower, together with currents in excess of 3A, traditional
linear regulators are simply too lossy to be used when
input supply is 5V or even in some cases with 3.3V
input supply. The IRU3037 together with dual N-channel
MOSFETs such as IRF7313, provide a low cost solution
for such applications. This device features an internal
200KHz oscillator (400KHz for "A" version), under-volt-
age lockout for both Vcc and Vc supplies, an external
programmable soft-start function as well as output un-
der-voltage detection that latches off the device when an
output short is detected.

Synchronous Controller in 8-Pin Package
Operating with single 5V or 12V supply voltage
Internal 200KHz Oscillator
(400KHz for IRU3037A)
Soft-Start Function
Fixed Frequency Voltage Mode
500mA Peak Output Drive Capability
Protects the output when control FET is shorted

PACKAGE ORDER INFORMATION

FEATURES

8-PIN SYNCHRONOUS PWM CONTROLLER

APPLICATIONS

DDR memory source sink Vtt application
Low cost on-board DC to DC such as 5V to 3.3V,
2.5V or 1.8V
Graphic Card
Hard Disk Drive

(癈)

DEVICE PACKAGE FREQUENCY

0 To 70

IRU3037CF 8-Pin Plastic TSSOP (F) 200KHz

0 To 70

IRU3037CS 8-Pin Plastic SOIC NB (S) 200KHz

0 To 70

IRU3037ACF 8-Pin Plastic TSSOP (F) 400KHz

0 To 70

IRU3037ACS 8-Pin Plastic SOIC NB (S) 400KHz

Data Sheet No. PD94173

Figure 1 - Typical application of IRU3037 or IRU3037A.

IRU3037

Vcc

HDrv

LDrv

Gnd

Comp

SS/SD

1uF

C4
1uF

0.1uF

2200pF

24k

Q1
1/2 of IRF7313

1/2 of IRF7313

R5
1.24K, 1%

249, 1%

D05022P-562, 5.6uH, 5.3A

1uH

C2
10TPB100M,

100uF, 55m

47uF

1.5V/5A

2x 6TPC150M,
150uF, 40m

12V

Rev. 2.8

05/10/04

IRU3037 / IRU3037A

www.irf.com

ABSOLUTE MAXIMUM RATINGS

Vcc Supply Voltage .................................................. 25V
Vc Supply Voltage ...................................................... 30V (not rated for inductive load)
Storage Temperature Range ...................................... -65癈 To 150癈
Operating Junction Temperature Range ..................... 0癈 To 125癈

CAUTION: Stresses above those listed in "Absolute Maximum Ratings" may cause permanent damage to the device.

PARAMETER

SYM TEST CONDITION

MIN TYP MAX

UNITS

Reference Voltage
Fb Voltage

Fb Voltage Line Regulation
UVLO
UVLO Threshold - Vcc
UVLO Hysteresis - Vcc
UVLO Threshold - Vc
UVLO Hysteresis - Vc
UVLO Threshold - Fb

UVLO Hysteresis - Fb
Supply Current
Vcc Dynamic Supply Current
Vc Dynamic Supply Current
Vcc Static Supply Current
Vc Static Supply Current
Soft-Start Section
Charge Current

IRU3037
IRU3037A
5<Vcc<12

Supply Ramping Up

Fb Ramping Down (IRU3037)
(IRU3037A)

Freq=200KHz, C

=1500pF

Freq=200KHz, C

=1500pF

SS=0V
SS=0V

SS=0V

1.225
0.784

4.0

3.1

0.4
0.3

2
2
1

0.5

-10

1.250
0.800

0.2

4.2

0.25

3.3
0.2
0.6
0.4
0.1

5
7

3.3

-20

1.275
0.816

0.35

4.4

3.5

0.8
0.5

4.5

-30

V
V
V
V
V

mA
mA
mA
mA

REG

UVLO Vcc

UVLO Vc

UVLO Fb

Dyn Icc

Dyn Ic

CCQ

=160

�

C/W

=124

�

C/W

Vcc

LDrv

Gnd

HDrv

Comp

SS/SD

Vcc

LDrv

Gnd

SS/SD

Comp

HDrv

ELECTRICAL SPECIFICATIONS

Unless otherwise specified, these specifications apply over Vcc=5V, Vc=12V and T

=0 to 70癈. Typical values refer

to T

=25癈. Low duty cycle pulse testing is used which keeps junction and case temperatures equal to the ambient

temperature.

PACKAGE INFORMATION

8-PIN PLASTIC TSSOP (F) 8-PIN PLASTIC SOIC (S)

IRU3037 / IRU3037A

Rev. 2.8
03/10/03

www.irf.com

PARAMETER

SYM TEST CONDITION

MIN TYP MAX

UNITS

PIN DESCRIPTIONS

This pin is connected directly to the output of the switching regulator via resistor divider to
provide feedback to the Error amplifier.

This pin provides biasing for the internal blocks of the IC as well as power for the low side
driver. A minimum of 1

F, high frequency capacitor must be connected from this pin to

ground to provide peak drive current capability.

Output driver for the synchronous power MOSFET.

This pin serves as the ground pin and must be connected directly to the ground plane. A
high frequency capacitor (0.1 to 1

F) must be connected from V5 and V12 pins to this pin

for noise free operation.

Output driver for the high side power MOSFET. Connect a diode, such as BAT54 or 1N4148,
from this pin to ground for the application when the inductor current goes negative (Source/
Sink), soft-start at no load and for the fast load transient from full load to no load.

This pin is connected to a voltage that must be at least 4V higher than the bus voltage of
the switcher (assuming 5V threshold MOSFET) and powers the high side output driver. A
minimum of 1

F, high frequency capacitor must be connected from this pin to ground to

provide peak drive current capability.

Compensation pin of the error amplifier. An external resistor and capacitor network is
typically connected from this pin to ground to provide loop compensation.

This pin provides soft-start for the switching regulator. An internal current source charges
an external capacitor that is connected from this pin to ground which ramps up the output
of the switching regulator, preventing it from overshooting as well as limiting the input
current. The converter can be shutdown by pulling this pin below 0.5V.

Error Amp
Fb Voltage Input Bias Current
Fb Voltage Input Bias Current
Transconductance
Oscillator
Frequency

Ramp-Amplitude Voltage
Output Drivers
Rise Time
Fall Time
Dead Band Time
Max Duty Cycle
Min Duty Cycle

SS=3V, Fb=1V
SS=0V, Fb=1V

IRU3037
IRU3037A

=1500pF

Fb=1V, Freq=200KHz
Fb=1.5V

410

180
345

1.225

50
85

-0.1

-64

600

200
400

1.25

50
50

150

830

220
440

1.275

100
100
250

mho

KHz

ns
ns
ns

%
%

PIN# PIN SYMBOL PIN DESCRIPTION

Vcc

LDrv

Gnd

HDrv

Comp

SS / SD

FB1

FB2

Freq

RAMP

OFF

Rev. 2.8

05/10/04

IRU3037 / IRU3037A

www.irf.com

BLOCK DIAGRAM

Figure 2 - Simplified block diagram of the IRU3037.

THEORY OF OPERATION

Introduction
The IRU3037 is a fixed frequency, voltage mode syn-
chronous controller and consists of a precision refer-
ence voltage, an error amplifier, an internal oscillator, a
PWM comparator, 0.5A peak gate driver, soft-start and
shutdown circuits (see Block Diagram).

The output voltage of the synchronous converter is set
and controlled by the output of the error amplifier; this is
the amplified error signal from the sensed output voltage
and the reference voltage.

This voltage is compared to a fixed frequency linear
sawtooth ramp and generates fixed frequency pulses of
variable duty-cycle, which drives the two N-channel ex-
ternal MOSFETs.The timing of the IC is provided through
an internal oscillator circuit which uses on-chip capaci-
tor to set the oscillation frequency to 200 KHz (400 KHz
for "A" version).

Soft-Start
The IRU3037 has a programmable soft-start to control
the output voltage rise and limit the current surge at the
start-up. To ensure correct start-up, the soft-start se-
quence initiates when the Vc and Vcc rise above their
threshold (3.3V and 4.2V respectively) and generates

the Power On Reset (POR) signal. Soft-start function
operates by sourcing an internal current to charge an
external capacitor to about 3V. Initially, the soft-start func-
tion clamps the E/A's output of the PWM converter. As
the charging voltage of the external capacitor ramps up,
the PWM signals increase from zero to the point the
feedback loop takes control.

Short-Circuit Protection
The outputs are protected against the short-circuit. The
IRU3037 protects the circuit for shorted output by sens-
ing the output voltage (through the external resistor di-
vider). The IRU3037 shuts down the PWM signals, when
the output voltage drops below 0.6V (0.4V for IRU3037A).

The IRU3037 also protects the output from over-voltaging
when the control FET is shorted. This is done by turning
on the sync FET with the maximum duty cycle.

Under-Voltage Lockout
The under-voltage lockout circuit assures that the
MOSFET driver outputs remain in the off state whenever
the supply voltage drops below set parameters. Lockout
occurs if Vc or Vcc fall below 3.3V and 4.2V respec-
tively. Normal operation resumes once Vc and Vcc rise
above the set values.

20uA

64uA Max

POR

Oscillator

Error Amp

Error Comp

Reset Dom

POR

0.5V

FbLo Comp

HDrv

Vcc

LDrv

Gnd

Vcc

4.0V

3.5V

0.2V

Bias

Generator

1.25V

POR

SS/SD

Fb 1

Comp 7

25K

1.25V

IRU3037 / IRU3037A

Rev. 2.8
03/10/03

www.irf.com

Figure 3 - Typical application of the IRU3037 for

programming the output voltage.

APPLICATION INFORMATION

Design Example:
The following example is a typical application for IRU3037,
the schematic is Figure 18 on page 14.

Output Voltage Programming
Output voltage is programmed by reference voltage and
external voltage divider. The Fb pin is the inverting input
of the error amplifier, which is internally referenced to
1.25V (0.8V for IRU3037A). The divider is ratioed to pro-
vide 1.25V at the Fb pin when the output is at its desired
value. The output voltage is defined by using the follow-
ing equation:

When an external resistor divider is connected to the
output as shown in Figure 3.

Equation (1) can be rewritten as:

Choose R

= 1K

This will result to

= 1.65K

If the high value feedback resistors are used, the input
bias current of the Fb pin could cause a slight increase
in output voltage. The output voltage set point can be
more accurate by using precision resistor.

Soft-Start Programming
The soft-start timing can be programmed by selecting
the soft start capacitance value. The start up time of the
converter can be calculated by using:

Where:
C

is the soft-start capacitor (

For a start-up time of 7.5ms, the soft-start capacitor will
be 0.1

F. Choose a ceramic capacitor at 0.1

Shutdown
The converter can be shutdown by pulling the soft-start
pin below 0.5V. The control MOSFET turns off and the
synchronous MOSFET turns on during shutdown.

Boost Supply Vc
To drive the high-side switch it is necessary to supply a
gate voltage at least 4V greater than the bus voltage.
This is achieved by using a charge pump configuration
as shown in Figure 18. The capacitor is charged up to
approximately twice the bus voltage. A capacitor in the
range of 0.1

F to 1

F is generally adequate for most

applications. In application, when a separate voltage
source is available the boost circuit can be avoided as
shown in Figure 1.

Input Capacitor Selection
The input filter capacitor should be based on how much
ripple the supply can tolerate on the DC input line. The
larger capacitor, the less ripple expected but consider
should be taken for the higher surge current during the
power-up. The IRU3037 provides the soft-start function
which controls and limits the current surge. The value of
the input capacitor can be calculated by the following
formula:

Where:
C

is the input capacitance (

is the input current (A)

t is the turn on time of the high-side switch (

V is the allowable peak to peak voltage ripple (V)

IRU3037

OUT

START

= 75

Css (ms) ---(2)

= 5V

OUT

= 3.3V

OUT

= 4A

OUT

= 100mV

= 200KHz

= R

- 1

OUT

REF

( )

OUT

= V

REF

1 +

---(1)

( )

= ---(3)

Rev. 2.8

05/10/04

IRU3037 / IRU3037A

www.irf.com

Assuming the following:

By using equation (3), C

= 193.3

For higher efficiency, low ESR capacitor is recommended.
Choose two 100

F capacitors.

The Sanyo TPB series PosCap capacitor 100

F, 10V

with 55m

ESR is a good choice.

Output Capacitor Selection
The criteria to select the output capacitor is normally
based on the value of the Effective Series Resistance
(ESR). In general, the output capacitor must have low
enough ESR to meet output ripple and load transient
requirements, yet have high enough ESR to satisfy sta-
bility requirements. The ESR of the output capacitor is
calculated by the following relationship:

The Sanyo TPC series, PosCap capacitor is a good
choice. The 6TPC150M 150

F, 6.3V has an ESR 40m

Selecting two of these capacitors in parallel, results to
an ESR of

20m

which achieves our low ESR goal.

The capacitor value must be high enough to absorb the
inductor's ripple current. The larger the value of capaci-
tor, the lower will be the output ripple voltage.

Inductor Selection
The inductor is selected based on output power, operat-
ing frequency and efficiency requirements. Low inductor
value causes large ripple current, resulting in the smaller
size, but poor efficiency and high output noise. Gener-
ally, the selection of inductor value can be reduced to
desired maximum ripple current in the inductor (

i). The

optimum point is usually found between 20% and 50%
ripple of the output current.

For the buck converter, the inductor value for desired
operating ripple current can be determined using the fol-
lowing relation:

i = 20%(I

), then the output inductor will be:

The Toko D124C series provides a range of inductors in
different values, low profile suitable for large currents,
10

H, 4.2A is a good choice for this application. This

will result to a ripple approximately 14% of output cur-
rent.

Power MOSFET Selection
The IRU3037 uses two N-Channel MOSFETs. The se-
lections criteria to meet power transfer requirements is
based on maximum drain-source voltage (V

DSS

), gate-

source drive voltage (V

), maximum output current, On-

resistance R

DS(ON)

and thermal management.

The MOSFET must have a maximum operating voltage
(V

DSS

) exceeding the maximum input voltage (V

The gate drive requirement is almost the same for both
MOSFETs. Logic-level transistor can be used and cau-
tion should be taken with devices at very low V

to pre-

vent undesired turn-on of the complementary MOSFET,
which results a shoot-through current.

The total power dissipation for MOSFETs includes con-
duction and switching losses. For the Buck converter
the average inductor current is equal to the DC load cur-
rent. The conduction loss is defined as:

The R

DS(ON)

temperature dependency should be consid-

ered for the worst case operation. This is typically given
in the MOSFET data sheet. Ensure that the conduction
losses and switching losses do not exceed the package
ratings or violate the overall thermal budget.

COND

(Upper Switch) = I

LOAD

DS(ON)

COND

(Lower Switch) = I

LOAD

DS(ON)

(1 - D)

= R

DS(ON)

Temperature Dependency

Where:

= Output Voltage Ripple

= Output Current

=100mV and

=4A

Results to ESR=25m

ESR

[

---(4)

Where:
V

= Maximum Input Voltage

OUT

= Output Voltage

i = Inductor Ripple Current

= Switching Frequency

t = Turn On Time

D = Duty Cycle

- V

OUT

= L

;

t = D

; D =

1
f

OUT

L = (V

- V

OUT

)

---(5)

OUT

L = 7

t = D

t = 3.3

1
f

= 2.93A

V = 1%(V

), Efficiency(

) = 90%

h 3

IRU3037 / IRU3037A

Rev. 2.8
03/10/03

www.irf.com

For this design, IRF7301 is a good choice. The device
provides low on-resistance in a compact SOIC 8-Pin
package.

The IRF7301 has the following data:

The total conduction losses will be:

The switching loss is more difficult to calculate, even
though the switching transition is well understood. The
reason is the effect of the parasitic components and
switching times during the switching procedures such
as turn-on / turnoff delays and rise and fall times. With a
linear approximation, the total switching loss can be ex-
pressed as:

The switching time waveform is shown in figure 4.

Figure 4 - Switching time waveforms.

From IRF7301 data sheet we obtain:

These values are taken under a certain condition test.
For more detail please refer to the IRF7301 data sheet.

By using equation (6), we can calculate the switching
losses.

Feedback Compensation
The IRU3037 is a voltage mode controller; the control
loop is a single voltage feedback path including error
amplifier and error comparator. To achieve fast transient
response and accurate output regulation, a compensa-
tion circuit is necessary. The goal of the compensation
network is to provide a closed loop transfer function with
the highest 0dB crossing frequency and adequate phase
margin (greater than 45

The output LC filter introduces a double pole, �40dB/
decade gain slope above its corner resonant frequency,
and a total phase lag of 180

(see Figure 5). The Reso-

nant frequency of the LC filter expressed as follows:

Figure 5 shows gain and phase of the LC filter. Since we
already have 180

phase shift just from the output filter,

the system risks being unstable.

Figure 5 - Gain and phase of LC filter.

The IRU3037's error amplifier is a differential-input
transconductance amplifier. The output is available for
DC gain control or AC phase compensation.

The E/A can be compensated with or without the use of
local feedback. When operated without local feedback
the transconductance properties of the E/A become evi-
dent and can be used to cancel one of the output filter
poles. This will be accomplished with a series RC circuit
from Comp pin to ground as shown in Figure 6.

DSS

= 20V

= 5.2A

DS(ON)

= 0.05

Where:
V

DS(OFF)

= Drain to Source Voltage at off time

= Rise Time

= Fall Time

T = Switching Period
I

LOAD

= Load Current

= 42ns

= 51ns

= 0.186W

CON(TOTAL)

CON

(Upper Switch)+P

CON

(Lower Switch)

CON(TOTAL)

= I

LOAD

DS(ON)

= 1.5 according to the IRF7301 data sheet for

150

C junction temperature

CON(TOTAL)

= 1.2W

= ---(7)

LOAD

---(6)

DS(OFF)

10%

90%

(ON)

(OFF)

Gain

0dB

Phase

-180

Frequency

-40dB/decade

Rev. 2.8

05/10/04

IRU3037 / IRU3037A

www.irf.com

Note that this method requires that the output capacitor
should have enough ESR to satisfy stability requirements.
In general the output capacitor's ESR generates a zero
typically at 5KHz to 50KHz which is essential for an
acceptable phase margin.

The ESR zero of the output capacitor expressed as fol-
lows:

Figure 6 - Compensation network without local

feedback and its asymptotic gain plot.

The transfer function (Ve / V

OUT

) is given by:

The (s) indicates that the transfer function varies as a
function of frequency. This configuration introduces a gain
and zero, expressed by:

The gain is determined by the voltage divider and E/A's
transconductance gain.

First select the desired zero-crossover frequency (Fo):

Use the following equation to calculate R

Where:
V

= Maximum Input Voltage

OSC

= Oscillator Ramp Voltage

Fo = Crossover Frequency
F

ESR

= Zero Frequency of the Output Capacitor

= Resonant Frequency of the Output Filter

and R

= Resistor Dividers for Output Voltage

Programming
g

= Error Amplifier Transconductance

This results to R

=104.4K

. Choose R

=105K

To cancel one of the LC filter poles, place the zero be-
fore the LC filter resonant frequency pole:

Using equations (11) and (13) to calculate C

, we get:

One more capacitor is sometimes added in parallel with
C

and R

. This introduces one more pole which is mainly

used to supress the switching noise. The additional pole
is given by:

The pole sets to one half of switching frequency which
results in the capacitor C

POLE:

For:
V

= 5V

OSC

= 1.25V

Fo = 30KHz
F

ESR

= 26.52KHz

= 2.9KHz

= 1K

= 1.65K

= 600

mho

= 698pF

Choose C

= 680pF

POLE

+ C

POLE

OUT

REF

E/A

H(s) dB

Frequency

Gain(dB)

Comp

ESR

= ---(8)

ESR

75%F

0.75

---(13)

For:
Lo = 10

Co = 300

= 2.17KHz

= 86.6K

Fo > F

ESR

and F

[

(1/5 ~ 1/10)

H(s) = g

3 3

---(9)

(

)

+ R

1 + sR

= ---(11)

|H(s)| = g

---(10)

---(12)

OSC

ESR

LC2

+ R

POLE

for F

IRU3037 / IRU3037A

Rev. 2.8
03/10/03

www.irf.com

For a general solution for unconditionally stability for any
type of output capacitors, in a wide range of ESR values
we should implement local feedback with a compensa-
tion network. The typically used compensation network
for voltage-mode controller is shown in Figure 7.

Figure 7 - Compensation network with local

feedback and its asymptotic gain plot.

In such configuration, the transfer function is given by:

The error amplifier gain is independent of the transcon-
ductance under the following condition:

By replacing Z

and Z

according to Figure 7, the trans-

former function can be expressed as:

As known, transconductance amplifier has high imped-
ance (current source) output, therefore, consider should
be taken when loading the E/A output. It may exceed its
source/sink output current capability, so that the ampli-
fier will not be able to swing its output voltage over the
necessary range.

The compensation network has three poles and two ze-
ros and they are expressed as follows:

Cross Over Frequency:

The stability requirement will be satisfied by placing the
poles and zeros of the compensation network according
to following design rules. The consideration has been
taken to satisfy condition (14) regarding transconduc-
tance error amplifier.

1) Select the crossover frequency:

Fo < F

ESR

and Fo

[

(1/10 ~ 1/6)

2) Select R

, so that R

3) Place first zero before LC's resonant frequency pole.

75% F

4) Place third pole at the half of the switching frequency.

> 50pF

If not, change R

selection.

5) Place R

in (15) and calculate C

1 -

1 +

OUT

Where:
V

= Maximum Input Voltage

OSC

= Oscillator Ramp Voltage

Lo = Output Inductor
Co = Total Output Capacitors

p 3

[

p 3

OSC

= 0

+ R

)

(

)

OUT

REF

E/A

Frequency

Gain(dB)

H(s) dB

Comp

>> 1 and

>>1 ---(14)

H(s)=

(1+sR

)

[1+sC

)]

)

1+sR

(1+sR

)

[

(

)]

= R

---(15)

OSC

Rev. 2.8

05/10/04

IRU3037 / IRU3037A

www.irf.com

6) Place second pole at the ESR zero.

= F

ESR

Check if R

If R

is too small, increase R

and start from step 2.

7) Place second zero around the resonant frequency.

= F

8) Use equation (1) to calculate R

These design rules will give a crossover frequency ap-
proximately one-tenth of the switching frequency. The
higher the band width, the potentially faster the load tran-
sient speed. The gain margin will be large enough to
provide high DC-regulation accuracy (typically -5dB to -
12dB). The phase margin should be greater than 45

for

overall stability.

IC Quiescent Power Dissipation
Power dissipation for IC controller is a function of ap-
plied voltage, gate driver loads and switching frequency.
The IC's maximum power dissipation occurs when the
IC operating with single 12V supply voltage (Vcc=12V
and Vc

24V) at 400KHz switching frequency and maxi-

mum gate loads.

Figures 9 and 10 show voltage vs. current, when the
gate drivers loaded with 470pF, 1150pF and 1540pF ca-
pacitors. The IC's power dissipation results to an exces-
sive temperature rise. This should be considered when
using IRU3037A for such application.

Layout Consideration
The layout is very important when designing high fre-
quency switching converters. Layout will affect noise
pickup and can cause a good design to perform with
less than expected results.

Start to place the power components, make all the con-
nection in the top layer with wide, copper filled areas.
The inductor, output capacitor and the MOSFET should
be close to each other as possible. This helps to reduce
the EMI radiated by the power traces due to the high
switching currents through them. Place input capacitor
directly to the drain of the high-side MOSFET, to reduce
the ESR replace the single input capacitor with two par-
allel units. The feedback part of the system should be
kept away from the inductor and other noise sources,
and be placed close to the IC. In multilayer PCB use
one layer as power ground plane and have a control cir-
cuit ground (analog ground), to which all signals are ref-
erenced. The goal is to localize the high current path to
a separate loop that does not interfere with the more
sensitive analog control function. These two grounds
must be connected together on the PC board layout at a
single point.

Figure 8 shows a suggested layout for the critical com-
ponents, based on the schematic on page 14.

Figure 8 - Suggested layout.

(Topside shown only)

p 3

= - R

p 3

REF

OUT

- V

REF

C
5

D
1

D
2

IRU3037

Vin

C2A, B

C7A, B

Analog Gnd

PGnd

Vout

PGnd

D
3

Single Point

Analog Gnd

Connect to

Power Ground plane

Analog Gnd

PGnd

Rev. 2.8

05/10/04

IRU3037 / IRU3037A

www.irf.com

TYPICAL PERFORMANCE CHARACTERISTICS

Figure 11 - Output Voltage

Figure 12 - Output Frequency

Figure 13 - Maximum Duty Cycle

I R U 3 0 3 7

O u t p u t V o l t a g e

1 . 2

1 . 2 2

1 . 2 4

1 . 2 6

1 . 2 8

1 . 3

-40癈

0 � C

+ 5 0 � C

+ 1 0 0 � C

+ 1 5 0 � C

Volts

O u t p u t V o l t a g e

S p e c M a x .

Spec Min.

M a x

I R U 3 0 3 7

O u t p u t F r e q u e n c y

1 6 0

1 7 0

1 8 0

1 9 0

2 0 0

2 1 0

2 2 0

2 3 0

2 4 0

-40癈

0 � C

+ 5 0 � C

+ 1 0 0 � C

+ 1 5 0 � C

Kilo Hertz

O s c i l l a t i o n F r e q u e n c y

S p e c M a x .

S p e c M i n .

M a x

IRU3037

Maximum Duty Cycle

80.0%

82.0%

84.0%

86.0%

88.0%

90.0%

92.0%

-40癈

-25癈

0癈

+25癈

+50癈

+75癈

+100癈 +125癈

+150癈

Percent Duty Cycle

Max Duty Cycle

Min

IRU3037 / IRU3037A

Rev. 2.8
03/10/03

www.irf.com

Figure 16 - Transconductance

Figure 15 - Output Frequency

Figure 14 - Output Voltage

Figure 17 - Rise Time and Fall Time

IRU3037 / IRU3037A

Rise Time / Fall Time

CL = 1500pF

-40癈

-25癈

0癈

+25癈

+50癈

+75癈

+100癈

nano Seconds

Rise Time

Fall time

IRU3037 / IRU3037A

Transconductance ( GM )

100

200

300

400

500

600

700

800

900

1000

-40癈

-25癈

0癈

+25癈

+50癈

+75癈

+100癈

micro Mho's

Positive load GM

Negative load GM

I R U 3 0 3 7 A

O u t p u t V o l t a g e

7 6 0

7 7 0

7 8 0

7 9 0

8 0 0

8 1 0

8 2 0

-40癈

- 2 5 � C

0 � C

+ 2 5 � C

+ 5 0 � C

+ 7 5 � C

+ 1 0 0 � C + 1 5 0 � C

milli Volts

O u t p u t V o l t a g e

S p e c M a x .

S p e c M i n .

M a x

Min

I R U 3 0 3 7 A

O u t p u t F r e q u e n c y

3 0 0

3 2 0

3 4 0

3 6 0

3 8 0

4 0 0

4 2 0

4 4 0

4 6 0

- 4 0 � C

-25癈

0癈

+ 2 5 � C

+ 5 0 � C

+ 7 5 � C

+ 1 0 0 � C + 1 5 0 � C

Kilo Hertz

O s c i l l a t i o n F r e q u e n c y

S p e c M a x .

S p e c M i n .

M a x

Min

TYPICAL PERFORMANCE CHARACTERISTICS

IRU3037 / IRU3037A

Rev. 2.8
03/10/03

www.irf.com

Figure 19 - Typical application of IRU3037 or IRU3037A in an on-board DC-DC converter providing the Core,

GTL+, and Clock supplies for the Pentium II microprocessor.

TYPICAL APPLICATION

Dual Supply, 5V Bus and 12V Bias Input

IRU3037

Vcc

HDrv

LDrv

Gnd

Comp

SS/SD

Vcc

HDrv

LDrv

Gnd

Comp

12V

IRU1206-18

2.5V/2A

1.8V/1A

47uF

1uH

10TPB100M, 100uF,

55m

, 1.5A rms

1/2 of IRF7752

CTX5-2P, 3.5uH @ 2.5A

1/2 of IRF7752

1K, 1%

1uF

0.1uF

2200pF

14K

C10

1uF

C11

1uF

C13

0.1uF

C14

2200pF

14K

1K, 1%

1.65K, 1%

1/2 of IRF7752

CTX5-1P, 3.4uH @ 2A

1/2 of IRF7752

10TPB100M, 100uF,

55m

, 1.5A rms

3.3V/1.8A

CTX5-2P, 3.5uH @ 2.5A

CTX10-5P, 5.7uH @ 2.5A

6TPB150M, 150uF, 55m

(Qty 2)

6TPB150M, 150uF, 55m

(Qty 1)

6TPB150M, 150uF, 55m

C12
6TPB150M, 150uF, 55m

IRU3037

SS/SD

IRU3037 / IRU3037A

Rev. 2.8
03/10/03

www.irf.com

DEMO-BOARD APPLICATION

5V or 12V to 3.3V @ 10A

Ref Desig

Description Value Qty

Part# Manuf

1
1
1
3
1
1
1
2
2
2
1
1
1
3
1
1
1
1

Q1
Q2
U1
D1, D2, D4
L1
L2
C1
C2A, C2B
C9B, C9C
C5, C6
C3
C4
C7
C8, C13, C19
R3
R4
R5
R6

MOSFET
MOSFET
Controller
Diode
Inductor
Inductor
Capacitor, Tantalum
Capacitor, Poscap
Capacitor, Poscap
Capacitor, Ceramic
Capacitor, Ceramic
Capacitor, Ceramic
Capacitor, Ceramic
Capacitor, Ceramic
Resistor
Resistor
Resistor
Resistor

IRF7457
IRF7460
IRU3037
LL4148
D03316P-102HC
D05022P-332HC
ECS-T1CD336R
16TPB47M
6TPC150M
ECJ-2VF1E104Z
ECJ-3YB1E105K
ECJ-2VB1H222K
ECJ-2VB2D471K
ECJ-2VF1C105Z

IR
IR
IR

Coilcraft
Coilcraft
Panasonic
Sanyo
Sanyo
Panasonic
Panasonic
Panasonic
Panasonic
Panasonic

Application Parts List

20V, 7m

, 15A

20V, 10m

, 12A

Synchronous PWM
Fast Switching
1

H, 10A

3.3

H, 12A

F, 16V

F, 16V, 70m

150

F, 6.3V, 40m

0.1

F, Y5V, 25V

F, X7R, 25V

2200pF, X7R, 50V
470pF, X7R
1

F, Y5V, 16V

20K, 5%
4.7

, 5%

1K, 1%
1.65K, 1%

Figure 21 - Demo-board application of IRU3037.

IRU3037

Vcc

HDrv

LDrv

Gnd

Comp

SS/SD

Gnd

5V or 12V

Q1
IRF7457

IRF7460

C2A

47uF

16V

C2B
47uF
16V

20K

C1
33uF

16V

D1
LL4148

LL4148

C6
0.1uF

3.3uH

C7
470pF

R4
4.7

C9B
150uF
6.3V

C13
1uF

1uH

C3
1uF

C19
1uF

1uF

0.1uF

2200pF

1.65K

LL4148

OUT

3.3V

@ 10A

R5
1K

C9C

150uF
6.3V

协谢械泻褌褉芯薪薪褘泄 泻芯屑锌芯薪械薪褌: IRU3037

协谢械泻褌褉芯薪薪褘泄泻芯屑锌芯薪械薪褌: IRU3037