Philips
Semiconductors
NE/SA5230
Low voltage operational amplifier
Product specification
1994 Aug 31
INTEGRATED CIRCUITS
Philips Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
2
1994 Aug 31
853-0942 13721
DESCRIPTION
The NE5230 is a very low voltage operational amplifier that can
perform with a voltage supply as low as 1.8V or as high as 15V. In
addition, split or single supplies can be used, and the output will
swing to ground when applying the latter. There is a bias adjusting
pin which controls the supply current required by the device and
thereby controls its power consumption. If the part is operated at
0.9V supply voltages, the current required is only 110
A when the
current control pin is left open. Even with this low power
consumption, the device obtains a typical unity gain bandwidth of
180kHz. When the bias adjusting pin is connected to the negative
supply, the unity gain bandwidth is typically 600kHz while the supply
current is increased to 600
A. In this mode, the part will supply full
power output beyond the audio range.
The NE5230 also has a unique input stage that allows the
common-mode input range to go above the positive and below the
negative supply voltages by 250mV. This provides for the largest
possible input voltages for low voltage applications. The part is also
internally-compensated to reduce external component count.
The NE5230 has a low input bias current of typically
40nA, and a
large open-loop gain of 125dB. These two specifications are
beneficial when using the device in transducer applications. The
large open-loop gain gives very accurate signal processing because
of the large "excess" loop gain in a closed-loop system.
The output stage is a class AB type that can swing to within 100mV
of the supply voltages for the largest dynamic range that is needed
in many applications. The NE5230 is ideal for portable audio
equipment and remote transducers because of its low power
consumption, unity gain bandwidth, and 30nV/
Hz noise
specification.
FEATURES
Works down to 1.8V supply voltages
Adjustable supply current
Low noise
Common-mode includes both rails
V
OUT
within 100mV of both rails
PIN CONFIGURATION
1
2
3
4
5
6
7
8
NC
IN
+IN
OUTPUT
BIAS ADJ.
VEE
VCC
NC
+
N, D, FE Packages
SP00250
Figure 1. Pin Configuration
APPLICATIONS
Portable precision instruments
Remote transducer amplifier
Portable audio equipment
Rail-to-rail comparators
Half-wave rectification without diodes
Remote temperature transducer with 4 to 20mA output
transmission
ORDERING INFORMATION
DESCRIPTION
TEMPERATURE RANGE
ORDER CODE
DWG #
8-Pin Plastic Small Outline (SO) Package
0 to +70
C
NE5230D
SOT96-1
8-Pin Plastic Dual In-Line Package (DIP)
0 to +70
C
NE5230N
SOT97-1
8-Pin Plastic Small Outline (SO) Package
-40
C to +85
C
SA5230D
SOT96-1
8-Pin Ceramic Dual In-Line Package (CERDIP)
-40
C to +85
C
SA5230FE
0580A
8-Pin Plastic Dual In-Line Package (DIP)
-40
C to +85
C
SA5230N
SOT97-1
Philips Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
1994 Aug 31
3
ABSOLUTE MAXIMUM RATINGS
SYMBOL
PARAMETER
RATING
UNIT
V
CC
Single supply voltage
18
V
V
S
Dual supply voltage
9
V
V
IN
Input voltage
1
9 (18)
V
Differential input voltage
1
V
S
V
V
CM
Common-mode voltage (positive)
V
CC
+0.5
V
V
CM
Common-mode voltage (negative)
V
EE
-0.5
V
P
D
Power dissipation
2
500
mW
T
J
Operating junction temperature
2
150
C
80Output short-circuit duration to either power supply pin
2, 3
Indefinite
s
T
STG
Storage temperature
-65 to 150
C
T
SOLD
Lead soldering temperature (10sec max)
300
C
NOTES:
1. Can exceed the supply voltages when V
S
7.5V (15V).
2. The maximum operating junction temperature is 150
C. At elevated temperatures, devices must be derated according to the package ther-
mal resistance and device mounting conditions. Derate above 25
C at the following rates:
FE package at 6.7mW/
C
N package at 9.5mW/
C
D package at 6.25mW/
C
3. Momentary shorts to either supply are permitted in accordance to transient thermal impedance limitations determined by the package and
device mounting conditions.
RECOMMENDED OPERATING CONDITIONS
PARAMETER
RATING
UNIT
Single supply voltage
1.8 to 15
V
Dual supply voltage
0.9 to
7.5
V
Common-mode voltage (positive)
V
CC
+0.25
V
Common-mode voltage (negative)
V
EE
-0.25
V
Temperature
NE grade
0 to 70
C
SA grade
-40 to 85
C
Philips Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
1994 Aug 31
4
DC AND AC ELECTRICAL CHARACTERISTICS
Unless otherwise specified,
0.9V
V
S
+7.5V or equivalent single supply, R
L
=10k
, full input common-mode range, over full operating
temperature range.
SYMBOL
PARAMETER
TEST CONDITIONS
BIAS
NE/SA5230
UNIT
SYMBOL
PARAMETER
TEST CONDITIONS
BIAS
Min
Typ
Max
UNIT
V
OS
Offset voltage
T
A
=25
C
Any
0.4
3
mV
V
OS
Offset voltage
T
A
=25
C
Any
3
4
mV
V
OS
Drift
Any
2
5
V/
C
T
A
=25
C
High
3
50
I
OS
Offset current
T
A
=25
C
Low
3
30
nA
I
OS
Offset current
High
100
nA
Low
60
I
OS
Drift
High
0.5
1.4
nA/
C
I
OS
Drift
Low
0.3
1.4
nA/
C
T
A
=25
C
High
40
150
I
Bias current
T
A
=25
C
Low
20
60
nA
I
B
Bias current
High
200
nA
Low
150
I
Drift
High
2
4
nA/
C
I
B
Drift
Low
2
4
nA/
C
T
A
=25
C
Low
110
160
V
S
=
0 9V
T
A
=25
C
High
600
750
A
V
S
=
0.9V
Low
250
A
I
S
Supply current
High
800
I
S
Supply current
T
A
=25
C
Low
320
550
V
S
=
7 5V
T
A
=25
C
High
1.1
1.6
A
V
S
=
7.5V
Low
600
A
High
1.7
V
C
Common mode input range
V
OS
6mV, T
A
=25
C
Any
V
-0.25
V
+
+0.25
V
V
CM
Common-mode input range
Any
V
V
+
V
CMRR
Common-mode rejection ratio
V
S
=
7.5V
R
S
=10k
, V
CM
=
7.5V,
T
A
=25
C
Any
85
95
dB
j
S
R
S
=10k
, V
CM
=
7.5V
Any
80
T
A
=25
C
High
90
105
PSRR
Power supply rejection ratio
T
A
=25
C
Low
85
95
dB
PSRR
Power supply rejection ratio
High
75
dB
Low
80
source
V
S
=
7.5V
Any
4
10
sink
V
S
=
7.5V
Any
5
15
source
V
S
=
7.5V
Any
1
5
I
Load current
sink
V
S
=
7.5V
Any
2
6
mA
I
L
Load current
source
V
S
=
0.9V, T
A
=25
C
High
4
6
mA
sink
V
S
=
0.9V, T
A
=25
C
High
5
7
source
V
S
=
7.5V, T
A
=25
C
High
16
sink
V
S
=
7.5V, T
A
=25
C
High
32
R
L
=10k
, T
A
=25
C
High
120
2000
V/mV
A
O
Large signal open loop gain
V
S
=
7 5V
R
L
=10k
, T
A
=25
C
Low
60
750
A
VOL
Large-signal open-loop gain
V
S
=
7.5V
High
100
V/mV
Low
50
Philips Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
1994 Aug 31
5
DC AND AC ELECTRICAL CHARACTERISTICS
(Continued)
SYMBOL
PARAMETER
TEST CONDITIONS
BIAS
NE/SA5230
UNIT
SYMBOL
PARAMETER
TEST CONDITIONS
BIAS
Min
Typ
Max
UNIT
T
A
=25
C +SW
Any
750
800
V
S
=
0 9V
T
A
=25
C -SW
Any
750
800
mV
V
S
=
0.9V
+SW
Any
700
mV
V
O
Output voltage swing
-SW
Any
700
V
OUT
Output voltage swing
T
A
=25
C +SW
Any
7.30
7.35
V
S
=
7.5V
T
A
=25
C -SW
Any
-7.32
-7.35
V
+SW
Any
7.25
7.30
V
-SW
Any
-7.30
-7.35
SR
Slew rate
T
A
=25
C
High
0.25
V/
s
SR
Slew rate
T
A
=25
C
Low
0.09
V/
s
BW
Inverting unity gain bandwidth
C
L
=100pF, T
A
=25
C
High
0.6
MHz
BW
Inverting unity gain bandwidth
C
L
=100pF, T
A
=25
C
Low
0.25
MHz
M
Phase margin
C
L
=100pF, T
A
=25
C
Any
70
Deg.
t
S
Settling time
C
L
=100pF, 0.1%
High
2
s
t
S
Settling time
C
L
=100pF, 0.1%
Low
5
s
V
Input noise
R
S
=0
, f=1kHz
High
30
nV/
Hz
V
INN
Input noise
R
S
=0
, f=1kHz
Low
60
nV/
Hz
THD
Total Harmonic Distortion
V
S
=
7.5V
A
V
=1, V
IN
=500mV, f=1kHz
High
0.003
%
THD
Total Harmonic Distortion
V
S
=
0.9V
A
V
=1, V
IN
=500mV, f=1kHz
High
0.002
%
Philips Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
1994 Aug 31
6
THEORY OF OPERATION
Input Stage
Operational amplifiers which are able to function at minimum supply
voltages should have input and output stage swings capable of
reaching both supply voltages within a few millivolts in order to
achieve ease of quiescent biasing and to have maximum
input/output signal handling capability. The input stage of the
NE5230 has a common-mode voltage range that not only includes
the entire supply voltage range, but also allows either supply to be
exceeded by 250mV without increasing the input offset voltage by
more than 6mV. This is unequalled by any other operational
amplifier today.
In order to accomplish the feat of rail-to-rail input common-mode
range, two emitter-coupled differential pairs are placed in parallel so
that the common-mode voltage of one can reach the positive supply
rail and the other can reach the negative supply rail. The simplified
schematic of Figure 2 shows how the complementary
emitter-coupler transistors are configured to form the basic input
stage cell. Common-mode input signal voltages in the range from
0.8V above V
EE
to V
CC
are handled completely by the NPN pair, Q3
and Q4, while common-mode input signal voltages in the range of
V
EE
to 0.8V above V
EE
are processed only by the PNP pair, Q1 and
Q2. The intermediate range of input voltages requires that both the
NPN and PNP pairs are operating. The collector currents of the
input transistors are summed by the current combiner circuit
composed of transistors Q8 through Q11 into one output current.
Transistor Q8 is connected as a diode to ensure that the outputs of
Q2 and Q4 are properly subtracted from those of Q1 and Q3.
The input stage was designed to overcome two important problems
for rail-to-rail capability. As the common-mode voltage moves from
the range where only the NPN pair was operating to where both of
the input pairs were operating, the effective transconductance would
change by a factor of two. Frequency compensation for the ranges
where one input pair was operating would, of course, not be optimal
for the range where both pairs were operating. Secondly, fast
changes in the common-mode voltage would abruptly saturate and
restore the emitter current sources, causing transient distortion.
These problems were overcome by assuring that only the input
transistor pair which is able to function properly is active. The NPN
pair is normally activated by the current source I
B1
through Q5 and
the current mirror Q6 and Q7, assuming the PNP pair is
non-conducting. When the common-mode input voltage passes
below the reference voltage, V
B1
=0.8V at the base of Q5, the
emitter current is gradually steered toward the PNP pair, away from
the NPN pair. The transfer of the emitter currents between the
complementary input pairs occurs in a voltage range of about
120mV around the reference voltage V
B1
. In this way the sum of the
emitter currents for each of the NPN and PNP transistor pairs is kept
constant; this ensures that the transconductance of the parallel
combination will be constant, since the transconductance of bipolar
transistors is proportional to their emitter currents.
An essential requirement of this kind of input stage is to minimize
the changes in input offset voltage between that of the NPN and
PNP transistor pair which occurs when the input common-mode
voltage crosses the internal reference voltage, V
B1
. Careful circuit
layout with a cross-coupled quad for each input pair has yielded a
typical input offset voltage of less than 0.3mV and a change in the
input offset voltage of less than 0.1mV.
V
V
R10
R11
R8
R9
Q3
Q1
Q2
Q4
Q10
Q11
Q5
Q6
Q7
Q8
Q9
VEE
VCC
IOUT
Vb2
+
+
Vb1
VIN
VIN+
Ib1
SL00251
Figure 2. Input Stage
Philips Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
1994 Aug 31
7
Output Stage
Processing output voltage swings that nominally reach to less than
100mV of either supply voltage can only be achieved by a pair of
complementary common-emitter connected transistors. Normally,
such a configuration causes complex feed-forward signal paths that
develop by combining biasing and driving which can be found in
previous low supply voltage designs. The unique output stage of the
NE5230 separates the functions of driving and biasing, as shown in
the simplified schematic of Figure 3, and has the advantage of a
shorter signal path which leads to increasing the effective
bandwidth.
This output stage consists of two parts: the Darlington output
transistors and the class AB control regulator. The output transistor
Q3 connected with the Darlington transistors Q4 and Q5 can source
up to 10mA to an output load. The output of NPN Darlington
connected transistors Q1 and Q2 together are able to sink an output
current of 10mA. Accurate and efficient class AB control is
necessary to insure that none of the output transistors are ever
completely cut off. This is accomplished by the differential amplifier
(formed by Q8 and Q9) which controls the biasing of the output
transistors. The differential amplifier compares the summed voltages
across two diodes, D1 and D2, at the base of Q8 with the summed
voltages across the base-emitter diodes of the output transistors Q1
and Q3. The base-emitter voltage of Q3 is converted into a current
by Q6 and R6 and reconverted into a voltage across the
base-emitter diode of Q7 and R7. The summed voltage across the
base-emitter diodes of the output transistors Q3 and Q1 is
proportional to the logarithm of the product of the push and pull
currents I
OP
and I
ON
, respectively. The combined voltages across
diodes D1 and D2 are proportional to the logarithm of the square of
the reference current I
B1
. When the diode characteristics and
temperatures of the pairs Q1, D1 and Q3, Q2 are equal, the relation
I
OP
I
ON
=I
B1
I
B1
is satisfied.
Separating the functions of biasing and driving prevents the driving
signals from becoming delayed by the biasing circuit. The output
Darlington transistors are directly accessible for in-phase driving
signals on the bases of Q5 and Q2. This is very important for simple
high-frequency compensation. The output transistors can be
high-frequency compensated by Miller capacitors CM1A and CM1B
connected from the collectors to the bases of the output Darlington
transistors.
A general-purpose op amp of this type must have enough open-loop
gain for applications when the output is driving a low resistance
load. The NE5230 accomplishes this by inserting an intermediate
common-emitter stage between the input and output stages. The
three stages provide a very large gain, but the op amp now has
three natural dominant poles -- one at the output of each
common-emitter stage. Frequency compensation is implemented
with a simple scheme of nested, pole-splitting Miller integrators. The
Miller capacitors CM1A and CM1B are the first part of the nested
structure, and provide compensation for the output and intermediate
stages. A second pair of Miller integrators provide pole-splitting
compensation for the pole from the input stage and the pole
resulting from the compensated combination of poles from the
intermediate and output stages. The result is a stable,
internally-compensated op amp with a phase margin of 70 degrees.
Q5
Ib1
Ib2
Ib3
Ib5
Ib4
Q4
Q6
Q8
Q9
D1
Q7
Q3
D2
Q1
Q2
R7
R6
CM1B
CM1A
Vb2
Vb5
VCC
VEE
VOUT
IOP
ION
SL00252
Figure 3. Output Stage
Philips Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
1994 Aug 31
8
THERMAL CONSIDERATIONS
When using the NE5230, the internal power dissipation capabilities
of each package should be considered. Philips Semiconductors
does not recommend operation at die temperatures above 110
C in
the SO package because of its inherently smaller package mass.
Die temperatures of 150
C can be tolerated in all the other
packages. With this in mind, the following equation can be used to
estimate the die temperature:
T
J
= T
A
+ (P
D
JA
) (1)
Where
T
A
5
AmbientTemperature
T
J
+
Die Temperature
P
D
5
Power Dissipation
+
(I
CC
x V
CC
)
q
JA
5
Packagethermalresistance
+
270
o
C W for SO
*
8 in PC
board mounting
See the packaging section for information regarding other methods
of mounting.
JA
=100
C/W for the plastic DIP;
JA
=110
C/W for the ceramic DIP.
The maximum supply voltage for the part is 15V and the typical
supply current is 1.1mA (1.6mA max). For operation at supply
voltages other than the maximum, see the data sheet for I
CC
versus
V
CC
curves. The supply current is somewhat proportional to
temperature and varies no more than 100
A between 25
C and
either temperature extreme.
Operation at higher junction temperatures than that recommended is
possible but will result in lower MTBF (Mean Time Between
Failures). This should be considered before operating beyond
recommended die temperature because of the overall reliability
degradation.
DESIGN TECHNIQUES AND APPLICATIONS
The NE5230 is a very user-friendly amplifier for an engineer to
design into any type of system. The supply current adjust pin (Pin 5)
can be left open or tied through a pot or fixed resistor to the most
negative supply (i.e., ground for single supply or to the negative
supply for split supplies). The minimum supply current is achieved
by leaving this pin open. In this state it will also decrease the
bandwidth and slew rate. When tied directly to the most negative
supply, the device has full bandwidth, slew rate and I
CC
. The
programming of the current-control pin depends on the trade-offs
which can be made in the designer's application. The graph in
Figure 4 will help by showing bandwidth versus I
CC
. As can be seen,
the supply current can be varied anywhere over the range of 100
A
to 600
A for a supply voltage of 1.8V. An external resistor can be
inserted between the current control pin and the most negative
supply. The resistor can be selected between 1
to 100k
to
provide any required supply current over the indicated range. In
addition, a small varying voltage on the bias current control pin could
be used for such exotic things as changing the gain-bandwidth for
voltage controlled low pass filters or amplitude modulation.
Furthermore, control over the slew rate and the rise time of the
amplifier can be obtained in the same manner. This control over the
slew rate also changes the settling time and overshoot in pulse
response applications. The settling time to 0.1% changes from 5
s
at low bias to 2
s at high bias. The supply current control can also
be utilized for wave-shaping applications such as for pulse or
triangular waveforms. The gain-bandwidth can be varied from
between 250kHz at low bias to 600kHz at high bias current. The
slew rate range is 0.08V/
s at low bias and 0.25V/
s at high bias.
POWER
SUPPL
Y
CURRENT (
A)
a. Unity Gain Bandwidth vs Power Supply Current for
V
CC
=
0.9V
b. I
CC
Current vs Bias Current Adjusting Resistor for
Several Supply Voltages
800
700
600
500
400
300
200
100
100
200
300
400 500 600 700
UNITY GAIN BANDWIDTH (kHz)
TA = 25
C
VCC = 15V
VCC = 12V
VCC = 9V
VCC = 6V
VCC = 3V
VCC = 2V
VCC = 1.8V
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0.0
I CURRENT (mA) CC
100
101
102
103
104
105
RADJ (
)
SL00253
Figure 4.
The full output power bandwidth range for V
CC
equals 2V, is above
40kHz for the maximum bias current setting and greater than 10kHz
at the minimum bias current setting.
If extremely low signal distortion (<0.05%) is required at low supply
voltages, exclude the common-mode crossover point (V
B1
) from the
common-mode signal range. This can be accomplished by proper
bias selection or by using an inverting amplifier configuration.
Most single supply designs necessitate that the inputs to the op amp
be biased between V
CC
and ground. This is to assure that the input
signal swing is within the working common-mode range of the
amplifier. This leads to another helpful and unique property of the
NE5230 that other CMOS and bipolar low voltage parts cannot
achieve. It is the simple fact that the input common-mode voltage
can go beyond either the positive or negative supply voltages. This
benefit is made very clear in a non-inverting voltage-follower
configuration. This is shown in Figure 5 where the input sine wave
allows an undistorted output sine wave which will swing less than
100mV of either supply voltage. Many competitive parts will show
severe clipping caused by input common-mode limitations. The
NE5230 in this configuration offers more freedom for quiescent
biasing of the inputs close to the positive supply rail where similar op
amps would not allow signal processing.
Philips Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
1994 Aug 31
9
V+
V
+
NE5230
OTHER
PARTS
V+
V
V+
V
V+
V
SL00254
Figure 5. In a Non-Inverting Voltage-Follower Configuration,
the NE5230 will Give Full Rail-to-Rail Swing. Other Low Voltage
Amplifiers will not Because they are Limited by their Input
Common-Mode Range and Output Swing Capability.
There are not as many considerations when designing with the
NE5230 as with other devices. Since the NE5230 is
internally-compensated and has a unity gain-bandwidth of 600kHz,
board layout is not so stringent as for very high frequency devices
such as the NE5205. The output capability of the NE5230 allows it
to drive relatively high capacitive loads and small resistive loads.
The power supply pins should be decoupled with a low-pass RC
network as close to the supply pins as possible to eliminate 60Hz
and other external power line noise, although the power supply
rejection ratio (PSRR) for the part is very high. The pinout for the
NE5230 is the same as the standard single op amp pinout with the
exception of the bias current adjusting pin.
REMOTE TRANSDUCER WITH CURRENT
TRANSMISSION
There are many ways to transmit information along two wires, but
current transmission is the most beneficial when the sensing of
remote signals is the aim. It is further enhanced in the form of 4 to
20mA information which is used in many control-type systems. This
method of transmission provides immunity from line voltage drops,
large load resistance variations, and voltage noise pickup. The zero
reference of 4mA not only can show if there is a break in the line
when no current is flowing, but also can power the transducer at the
remote location. Usually the transducer itself is not equipped to
provide for the current transmission. The unique features of the
NE5230 can provide high output current capability coupled with low
power consumption. It can be remotely connected to the transducer
to create a current loop with minimal external components. The
circuit for this is shown in Figure 6. Here, the part is configured as a
voltage-to-current, or transconductance amplifier. This is a novel
circuit that takes advantage of the NE5230's large open-loop gain.
In AC applications, the load current will decrease as the open-loop
gain rolls off in magnitude. The low offset voltage and current
sinking capabilities of the NE5230 must also be considered in this
application.
a. The NE5230 as a Remote Transducer Transconductance
Amp With 4-20mA Current Transmission Output Capability
b. The Same Type of Circuit as Figure 5a, but for Sourcing
Current to the Load
T
R
A
N
S
D
U
C
E
R
V
REMOTE
POWER
SUPPLY
NE5230
VCC
VEE
VIN
IOUT
3
2
4
5
6
7
+
RC
RL
+
NOTES:
1. IOUT = VIN/RC
2. RL MAX
V
REMOTE
*
1.8V
*
V
INMAX
I
OUT
For RC = 1
I
OUT
4mA
20mA
V
IN
4mV
20mV
VCC
NE5230
3
2
4
5
6
7
+
VEE
+ IOUT
+
VIN
RC
RL
VCC
+
SL00255
Figure 6.
Philips Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
1994 Aug 31
10
The NE5230 circuit shown in Figure 6 is a pseudo transistor
configuration. The inverting input is equivalent to the "base," the
point where V
EE
and the non-inverting input meet is the "emitter,"
and the connection after the output diode meets the V
CC
pin is the
collector. The output diode is essential to keep the output from
saturating in this configuration. From here it can be seen that the
base and emitter form a voltage-follower and the voltage present at
R
C
must equal the input voltage present at the inverting input. Also,
the emitter and collector form a current-follower and the current
flowing through R
C
is equivalent to the current through R
L
and the
amplifier. This sets up the current loop. Therefore, the following
equation can be formulated for the working current transmission line.
The load current is:
I
L
=V
IN
/R
C
(2)
and proportional to the input voltage for a set R
C
. Also, the current is
constant no matter what load resistance is used while within the
operating bandwidth range of the op amp. When the NE5230's
supply voltage falls past a certain point, the current cannot remain
constant. This is the "voltage compliance" and is very good for this
application because of the near rail output voltage. The equation
that determines the voltage compliance as well as the largest
possible load resistor for the NE5230 is as follows:
R
L max
=[V
remote supply
)-V
CC min
- V
IN max
]/I
L
(3)
Where V
CC min
is the worst-case power supply voltage
(approximately 1.8V) that will still keep the part operational. As an
example, when using a 15V remote power supply, a current sensing
resistor of 1
, and an input voltage (V
IN
) of 20mV, the output current
(I
L
) is 20mA. Furthermore, a load resistance of zero to
approximately 650
can be inserted in the loop without any change
in current when the bias current-control pin is tied to the negative
supply pin. The voltage drop across the load and line resistance will
not affect the NE5230 because it will operate down to 1.8V. With a
15V remote supply, the voltage available at the amplifier is still
enough to power it with the maximum 20mA output into the 650
load.
What this means is that several instruments, such as a chart
recorder, a meter, or a controller, as well as a long cable, can be
connected in series on the loop and still obtain accurate readings if
the total resistance does not exceed 650
. Furthermore, any
variation of resistance in this range will not change the output
current.
Any voltage output type transducer can be used, but one that does
not need external DC voltage or current excitation to limit the
maximum possible load resistance is preferable. Even this problem
can be surmounted if the supply power needed by the transducer is
compatible with the NE5230. The power goes up the line to the
transducer and amplifier while the transducer signal is sent back via
the current output of the NE5230 transconductance configuration.
The voltage range on the input can be changed for transducers that
produce a large output by simply increasing the current sense
resistor to get the corresponding 4 to 20mA output current. If a very
long line is used which causes high line resistance, a current
repeater could be inserted into the line. The same configuration of
Figure 6 can be used with exception of a resistor across the input
and line ground to convert the current back to voltage. Again, the
current sensing resistor will set up the transconductance and the
part will receive power from the line.
TEMPERATURE TRANSDUCER
A variation on the previous circuit makes use of the supply current
control pin. The voltage present at this pin is proportional to absolute
temperature (PTAT) because it is produced by the amplifier bias
current through an internal resistor divider in a PTAT cell. If the
control pin is connected to the input pin, the NE5230 itself can be
used as a temperature transducer. If the center tap of a resistive pot
is connected to the control pin with one side to ground and the other
to the inverting input, the voltage can be changed to give different
temperature versus output current conditions (see Figure 7). For
additional control, the output current is still proportional to the input
voltage differential divided by the current sense resistor.
When using the NE5230 as a temperature transducer, the thermal
considerations in the previous section must be kept in mind.
V
REMOTE
POWER
SUPPLY
NE5230
VCC
VEE
IOUT
3
2
4
5
6
7
+
RC
RL
+
NOTES:
1. IOUT = VIN/RC
2. RL MAX
V
REMOTE
*
1.8V
*
V
INMAX
I
OUT
For RC = 1
I
OUT
4mA
20mA
V
IN
4mV
20mV
10
200
SL00256
Figure 7. NE5230 Remote Temperature Transducer Utilizing
4-20mA Current Transmission. This Application Shows the use
of the Accessibility of the PTAT Cell in the Device to Make the
Part, Itself, a Transducer
Philips Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
1994 Aug 31
11
HALF-WAVE RECTIFIER WITH RAIL-TO-GROUND
OUTPUT SWING
Since the NE5230 input common-mode range includes both positive
and negative supply rails and the output can also swing to either
supply, achieving half-wave rectifier functions in either direction
becomes a simple task. All that is needed are two external resistors;
there is no need for diodes or matched resistors. Moreover, it can
have either positive- or negative-going outputs, depending on the
way the bias is arranged. This can be seen in Figure 8. Circuit (a) is
biased to ground, while circuit (b) is biased to the positive supply.
This rather unusual biasing does not cause any problems with the
NE5230 because of the unique internal saturation detectors
incorporated into the part to keep the PNP and NPN output
transistors out of "hard" saturation. It is therefore relatively quick to
recover from a saturated output condition. Furthermore, the device
does not have parasitic current draw when the output is biased to
either rail. This makes it possible to bias the NE5230 into
"saturation" and obtain half-wave rectification with good recovery.
The simplicity of biasing and the rail-to-ground half-sine wave swing
are unique to this device. The circuit gain can be changed by the
standard op amp gain equations for an inverting configuration.
It can be seen in these configurations that the op amp cannot
respond to one-half of the incoming waveform. It cannot respond
because the waveform forces the amplifier to swing the output
beyond either ground or the positive supply rail, depending on the
biasing, and, also, the output cannot disengage during this half
cycle. During the other half cycle, however, the amplifier achieves a
half-wave that can have a peak equal to the total supply voltage.
The photographs in Figure 9 show the effect of the different biasing
schemes, as well as the wide bandwidth (it works over the full audio
range), that the NE5230 can achieve in this configuration.
By adding another NE5230 in an inverting summer configuration at
the output of the half-wave rectifier, a full-wave can be realized. The
values for the input and feedback resistors must be chosen so that
each peak will have equal amplitudes. A table for calculating values
is included in Figure 10. The summing network combines the input
signal at the half-wave and adds it to double the half-wave's output,
resulting in the full-wave. The output waveform can be referenced to
the supply or ground, depending on the half-wave configuration.
Again, no diodes are needed to achieve the rectification.
This circuit could be used in conjunction with the remote transducer
to convert a received AC output signal into a DC level at the
full-wave output for meters or chart recorders that need DC levels.
VCC
a. Rail-to-Ground Output Swing Referenced to Ground
b. Negative-Going Output Referenced to V
CC
VIN
VCC
10
2
3
4
5
6
7
+
O
t
10
VCC
VOUT
t
VCC
10
3
2
4
5
6
7
10
VIN
VOUT
VCC
+
SL00257
Figure 8. Half-Wave Rectifier With Positive-Going Output Swings
Philips Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
1994 Aug 31
12
500mV/Div 200
S/Div
Biased to Ground
500mV/Div 20
S/Div
Biased to Ground
500mV/Div 20
S/Div
Biased to Positive Rail
SL00258
Figure 9. Performance Waveforms for the Circuits in Figure 8.
Good Response is Shown at 1and 10kHz for Both Circuits Under Full Swing With a 2V Supply
Philips Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
1994 Aug 31
13
HALF-WAVE
a
b
+VIN
VIN
a
b
+VIN
VIN
a
b
+VIN
+VB
FULL-WAVE
2a
2VIN
0
VEE
VEE
VCC
+VB
R1
3
2
4
5
6
7
R2
R3
R5
R4
2
3
4
5
6
7
+
+
A1
A2
INPUT
HALF-WAVE
OUTPUT
FULL-WAVE
OUTPUT
500mV
500mV
500mV
520
s
NOTES:
R2 = 2 R1
R4 = R5 = R3
+VB will vary output reference.
For single supply operation VEE can be grounded on A2.
SL00259
Figure 10. Adding an Inverting Summer to the Input and Output of the Half-Wave will Result in Full-Wave
CONCLUSION
The NE5230 is a versatile op amp in its own right. The part was
designed to give low voltage and low power operation without the
limitations of previously available amplifiers that had a multitude of
problems. The previous application examples are unique to this
amplifier and save the user money by excluding various passive
components that would have been needed if not for the NE5230's
special input and output stages.
The NE5230 has a combination of novel specifications which allows
the designer to implement it easily into existing low-supply voltage
designs and to enhance their performance. It also offers the
engineer the freedom to achieve greater amplifier system design
goals. The low input referenced noise voltage eases the restrictions
on designs where S/N ratios are important. The wide full-power
bandwidth and output load handling capability allow it to fit into
portable audio applications. The truly ample open-loop gain and low
power consumption easily lend themselves to the requirements of
remote transducer applications. The low, untrimmed typical offset
voltage and low offset currents help to reduce errors in signal
processing designs. The amplifier is well isolated from changes on
the supply lines by its typical power supply rejection ratio of 105dB.
REFERENCES
Johan H. Huijsing,
Multi-stage Amplifier with Capacitive Nesting for
Frequency Compensation, U.S. Patent Application Serial
No. 602.234, filed April 19, 1984.
Bob Blauschild,
Differential Amplifier with Rail-to-Rail Capability,
U.S. Patent Application Serial No. 525.181, filed August 23, 1983.
Operational Amplifiers - Characteristics and Applications,
Robert G. Irvine, Prentice-Hall, Inc., Englewood Cliffs, NJ 07632, 1981.
Transducer Interface Handbook - A Guide to Analog Signal
Conditioning, Edited by Daniel H. Sheingold, Analog Devices, Inc.,
Norwood, MA 02062, 1981.
Philips Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
August 31, 1994
15
SO8:
plastic small outline package; 8 leads; body width 3.9mm
SOT96-1
Philips Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
August 31, 1994
16
DIP8:
plastic dual in-line package; 8 leads (300 mil)
SOT97-1
Philips
Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
August
31, 1994
17
0580A
8-PIN (300 mils wide) CERAMIC DUAL IN-LINE (F) P
ACKAGE
NOTES:
1. Controlling dimension: Inches. Millimeters are
2. Dimension and tolerancing per ANSI Y14. 5M-1982.
3. "T", "D", and "E" are reference datums on the body
4. These dimensions measured with the leads
5. Pin numbers start with Pin #1 and continue
and include allowance for glass overrun and meniscus
on the seal line, and lid to base mismatch.
constrained to be perpendicular to plane T.
counterclockwise to Pin #8 when viewed
shown in parentheses.
from the top.
0.200 (5.08)
0.010 (0.254)
T
E D
0.023 (0.58)
0.015 (0.38)
0.165 (4.19)
0.125 (3.18)
0.165 (4.19)
0.175 (4.45)
0.145 (3.68)
0.320 (8.13)
0.290 (7.37)
(NOTE 4)
BSC
0.300 (7.62)
0.395 (10.03)
0.300 (7.62)
(NOTE 4)
0.015 (0.38)
0.010 (0.25)
0.035 (0.89)
0.020 (0.51)
D
PIN # 1
E
0.303 (7.70)
0.245 (6.22)
0.100 (2.54) BSC
0.408 (10.36)
0.376 (9.55)
0.055 (1.40)
0.030 (0.76)
T
SEATING
PLANE
0.070 (1.78)
0.050 (1.27)
0.055 (1.40)
0.030 (0.76)
8530580A
006688
Philips Semiconductors
Product specification
NE/SA5230
Low voltage operational amplifier
August 31, 1994
18
Philips Semiconductors and Philips Electronics North America Corporation reserve the right to make changes, without notice, in the products,
including circuits, standard cells, and/or software, described or contained herein in order to improve design and/or performance. Philips
Semiconductors assumes no responsibility or liability for the use of any of these products, conveys no license or title under any patent, copyright,
or mask work right to these products, and makes no representations or warranties that these products are free from patent, copyright, or mask
work right infringement, unless otherwise specified. Applications that are described herein for any of these products are for illustrative purposes
only. Philips Semiconductors makes no representation or warranty that such applications will be suitable for the specified use without further testing
or modification.
LIFE SUPPORT APPLICATIONS
Philips Semiconductors and Philips Electronics North America Corporation Products are not designed for use in life support appliances, devices,
or systems where malfunction of a Philips Semiconductors and Philips Electronics North America Corporation Product can reasonably be expected
to result in a personal injury. Philips Semiconductors and Philips Electronics North America Corporation customers using or selling Philips
Semiconductors and Philips Electronics North America Corporation Products for use in such applications do so at their own risk and agree to fully
indemnify Philips Semiconductors and Philips Electronics North America Corporation for any damages resulting from such improper use or sale.
This data sheet contains preliminary data, and supplementary data will be published at a later date. Philips
Semiconductors reserves the right to make changes at any time without notice in order to improve design
and supply the best possible product.
Philips Semiconductors and Philips Electronics North America Corporation
register eligible circuits under the Semiconductor Chip Protection Act.
Copyright Philips Electronics North America Corporation 1994
All rights reserved. Printed in U.S.A.
Philips Semiconductors
811 East Arques Avenue
P.O. Box 3409
Sunnyvale, California 940883409
Telephone 800-234-7381
DEFINITIONS
Data Sheet Identification
Product Status
Definition
Objective Specification
Preliminary Specification
Product Specification
Formative or in Design
Preproduction Product
Full Production
This data sheet contains the design target or goal specifications for product development. Specifications
may change in any manner without notice.
This data sheet contains Final Specifications. Philips Semiconductors reserves the right to make changes
at any time without notice, in order to improve design and supply the best possible product.
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